(19)
(11)EP 3 219 023 B1

(12)EUROPEAN PATENT SPECIFICATION

(45)Mention of the grant of the patent:
29.04.2020 Bulletin 2020/18

(21)Application number: 15858802.0

(22)Date of filing:  10.11.2015
(51)International Patent Classification (IPC): 
H04B 7/04(2017.01)
H04B 7/0456(2017.01)
H04B 17/12(2015.01)
H01Q 1/24(2006.01)
H04B 17/373(2015.01)
H01Q 21/06(2006.01)
H04N 7/06(2006.01)
H01Q 3/26(2006.01)
H04B 7/0452(2017.01)
H01Q 21/22(2006.01)
H04B 7/06(2006.01)
H04L 5/00(2006.01)
(86)International application number:
PCT/KR2015/012075
(87)International publication number:
WO 2016/076614 (19.05.2016 Gazette  2016/20)

(54)

2D ACTIVE ANTENNA ARRAY OPERATION FOR WIRELESS COMMUNICATION SYSTEMS

BETRIEB EINER 2D-AKTIVGRUPPENANTENNE FÜR DRAHTLOSKOMMUNIKATIONSSYSTEME

FONCTIONNEMENT D'UN RÉSEAU D'ANTENNES ACTIVES 2D POUR LES SYSTÈMES DE COMMUNICATION SANS FIL


(84)Designated Contracting States:
AL AT BE BG CH CY CZ DE DK EE ES FI FR GB GR HR HU IE IS IT LI LT LU LV MC MK MT NL NO PL PT RO RS SE SI SK SM TR

(30)Priority: 10.11.2014 US 201462077795 P
14.11.2014 US 201462080090 P
30.12.2014 US 201462098092 P
06.11.2015 US 201514935172

(43)Date of publication of application:
20.09.2017 Bulletin 2017/38

(73)Proprietor: Samsung Electronics Co., Ltd.
Gyeonggi-do 16677 (KR)

(72)Inventors:
  • YUAN, Jin
    Richardson, Texas 75082 (US)
  • LI, Yang
    Plano, Texas 75025 (US)
  • NAM, Young-Han
    Plano, Texas 75025 (US)
  • TZANIDIS, Ioannis
    Dallas, Texas 75243 (US)
  • XU, Gang
    Allen, Texas 75013 (US)
  • RAHMAN, Md. Saifur
    Richardson, Texas 75081 (US)
  • XIN, Yan
    Princeton, New Jersey 08540 (US)
  • MONROE, Robert
    Melissa, Texas 75454 (US)
  • ZHANG, Jianzhong
    Plano, Texas 75093 (US)
  • ONGGOSANUSI, Eko
    Allen, Texas 75013 (US)

(74)Representative: HGF Limited 
Saviour House 9 St. Saviourgate
York YO1 8NQ
York YO1 8NQ (GB)


(56)References cited: : 
EP-A2- 2 775 634
WO-A1-2013/169389
US-A1- 2011 243 272
US-A1- 2014 242 914
EP-A2- 2 775 634
WO-A1-2014/007512
US-A1- 2014 098 689
US-A1- 2014 242 914
  
      
    Note: Within nine months from the publication of the mention of the grant of the European patent, any person may give notice to the European Patent Office of opposition to the European patent granted. Notice of opposition shall be filed in a written reasoned statement. It shall not be deemed to have been filed until the opposition fee has been paid. (Art. 99(1) European Patent Convention).


    Description

    Technical Field



    [0001] The present application relates generally to wireless communication systems and, more specifically, to a 2 dimensional active antenna array operation for wireless communication systems.

    Background Art



    [0002] To meet the demand for wireless data traffic having increased since deployment of 4G communication systems, efforts have been made to develop an improved 5G or pre-5G communication system. Therefore, the 5G or pre-5G communication system is also called a 'Beyond 4G Network' or a 'Post LTE System'.

    [0003] The 5G communication system is considered to be implemented in higher frequency (mmWave) bands, e.g., 60GHz bands, so as to accomplish higher data rates. To decrease propagation loss of the radio waves and increase the transmission distance, the beamforming, massive multiple-input multiple-output (MIMO), Full Dimensional MIMO (FD-MIMO), array antenna, an analog beam forming, large scale antenna techniques are discussed in 5G communication systems.

    [0004] In addition, in 5G communication systems, development for system network improvement is under way based on advanced small cells, cloud Radio Access Networks (RANs), ultra-dense networks, device-to-device (D2D) communication, wireless backhaul, moving network, cooperative communication, Coordinated Multi-Points (CoMP), reception-end interference cancellation and the like.

    [0005] In the 5G system, Hybrid FSK and QAM Modulation (FQAM) and sliding window superposition coding (SWSC) as an advanced coding modulation (ACM), and filter bank multi carrier(FBMC), non-orthogonal multiple access(NOMA), and sparse code multiple access (SCMA) as an advanced access technology have been developed.

    [0006] A FD-MIMO system may support up to 64 antenna ports in a 2 dimensional (2D) array while providing enhanced performance. Therefore, the FD-MIMO system is considered as a key area in LTE standardization. The FD-MIMO system may provide enhanced system performance without requiring a very higher performance backhaul or large frequency resources compared to the CoMP and a carrier aggregation (CA) technique. However, there is a big challenge to accommodate a high-order multiuser MIMO (MU-MIMO) transmission and reception without complicating design and implementation of both base station and user equipment (UE) because the higher-order MU-MIMO refers to the use of a large number of antennas at the base station in order to transmit and receive spatially multiplexed signals to and from a large number of UEs.

    [0007] EP 2775 634 discloses a method for multi-input multi-output transmission of a base station in a wireless communication system. The method includes obtaining channel information of one or more terminals, classifying the one or more terminals into one or more classes and one or more groups dependent on the class based on the channel information, determining a group beamforming matrix for each of the one or more groups, performing group beamforming transmission on terminals belonging to each of the one or more groups based on the group beamforming matrix, obtaining single user-channel quality indicator (SU-CQI) information and interference signal information of each of the terminals belonging to each of the one or more groups, and scheduling the terminals based on the SU-CQI information and the interference signal information.

    [0008] US 2014/242914 discloses a method and apparatus for calibrating multiple antenna arrays.

    [0009] WO 2014/007512 discloses a method and device for reporting channel state information in a wireless communication system.

    Disclosure of Invention


    Solution to Problem



    [0010] Embodiments of the present disclosure provide a 2D active antenna array operation for wireless communication systems.

    [0011] In one embodiment, a method for operating a base station, BS, in a wireless communication system, the method comprising: receiving a plurality of signals including information for beamforming to a plurality of user equipments, UEs, using a full-dimensional multiple-input multiple-output, D-MIMO, beamforming scheme through at least one time resource and at least one frequency resource that are co-scheduled to the plurality of UEs; performing a calibration for generating a precoder based on the received plurality of signals, wherein the performing the calibration comprises: detecting a sample level mismatch for a plurality of common public radio interface, CPRI, connections associated with at least one antenna array; compensating for the detected sample level mismatch for the plurality of CPRI connections; detecting delay differences across a plurality of channels after the detected sample level mismatch is compensated; compensating for the detected delay differences across the plurality of channels; and adjusting phases of the plurality of channels to be in-phase after the detected delay differences are compensated; and performing the beamforming on signals to the plurality of UEs based on a precoder obtained from a result of the calibration.

    [0012] An apparatus for performing this method is also provided.

    [0013] The invention is defined by the independent claims. Preferred embodiments of the invention are stipulated in the dependent claims. While several embodiments and/or examples have been disclosed in this description, the subject matter for which protection is sought is strictly and solely limited to those embodiments and/or examples encompassed by the scope of the appended claims. Embodiments and/or examples mentioned in the description that do not fall under the scope of the claims are useful for understanding the invention.

    [0014] Before undertaking the DETAILED DESCRIPTION below, it may be advantageous to set forth definitions of certain words and phrases used throughout this patent document: the terms "include" and "comprise," as well as derivatives thereof, mean inclusion without limitation; the term "or," is inclusive, meaning and/or; the phrases "associated with" and "associated therewith," as well as derivatives thereof, may mean to include, be included within, interconnect with, contain, be contained within, connect to or with, couple to or with, be communicable with, cooperate with, interleave, juxtapose, be proximate to, be bound to or with, have, have a property of, or the like; and the term "controller" means any device, system or part thereof that controls at least one operation, such a device may be implemented in hardware, firmware or software, or some combination of at least two of the same. It should be noted that the functionality associated with any particular controller may be centralized or distributed, whether locally or remotely. Definitions for certain words and phrases are provided throughout this patent document, those of ordinary skill in the art should understand that in many, if not most instances, such definitions apply to prior, as well as future uses of such defined words and phrases.

    Brief Description of Drawings



    [0015] For a more complete understanding of the present disclosure and its advantages, reference is now made to the following description taken in conjunction with the accompanying drawings, in which like reference numerals represent like parts:

    FIG. 1 illustrates an example configuration of a full-dimensional multiple input multiple output (FD-MIMO) system according to an exemplary embodiment of the disclosure;

    FIG. 2 illustrates an example message flow of an FD-MIMO system according to an exemplary embodiment of the disclosure;

    FIG. 3 illustrates an example hardware (HW) configuration of an FD-MIMO system according to an exemplary embodiment of the disclosure;

    FIG. 4 illustrates an example unit diagram of an baseband processing for an FD-MIMO system according to an exemplary embodiment of the disclosure;

    FIG. 5 illustrates an example flowchart of a basic processing method for an FD-MIMO system according to an exemplary embodiment of the disclosure;

    FIG. 6 illustrates an example configuration of a 4 FD-MIMO antenna array architecture according to an exemplary embodiment of the disclosure;

    FIG. 7 illustrates an example configuration of an FD-MIMO 2 dimensional (D) antenna array according to an exemplary embodiment of the disclosure;

    FIG. 8 illustrates an example unit diagram of an FD-MIMO antenna array visualization according to an exemplary embodiment of the disclosure;

    FIG. 9 illustrates an example configuration of an FD-MIMO antenna array visualization according to an exemplary embodiment of the disclosure;

    FIG. 10 illustrates an example configuration of an FD-MIMO beam forming in a 2D large scale antenna array according to an exemplary embodiment of the disclosure;

    FIG. 11 illustrates an example performance result of an FD-MIMO 2D antenna array according to an exemplary embodiment of the disclosure;

    FIG. 12 illustrates an example unit diagram of an eNodeB (eNB) processing chain with a multi-user channel quality indication (MU-CQI) prediction according to an exemplary embodiment of the disclosure;

    FIG. 13 illustrates an example unit diagram of a multi-user channel quality indication (MU-CQI) prediction according to an exemplary embodiment of the disclosure;

    FIG. 14 illustrates an example simulation result of a single user CQI (SU-CQI) according to an exemplary embodiment of the disclosure;

    FIG. 15 illustrates an example configuration of a sounding reference signal (SRS) channel assignment according to an exemplary embodiment of the disclosure;

    FIG. 16 illustrates an example flowchart of an SRS based channel estimation and per resource unit (RB) precoder generation method according to an exemplary embodiment of the disclosure;

    FIG. 17 illustrates an example flowchart of a calibration method according to an exemplary embodiment of the disclosure;

    FIG. 18 illustrates an example unit diagram of a calibration circuit according to an exemplary embodiment of the disclosure;

    FIG. 19 illustrates an example unit diagram of a large-scale antenna system according to an exemplary embodiment of the disclosure;

    FIG. 20 illustrates operations of a base station according to an exemplary embodiment of the disclosure; and

    FIG. 21 illustrates operations of a user equipment according to an exemplary embodiment of the disclosure.


    Best Mode for Carrying out the Invention



    [0016] FIG. 1 through FIG. 21, discussed below, and the various embodiments used to describe the principles of the present disclosure in this patent document are by way of illustration only and should not be construed in any way to limit the scope of the disclosure. Those skilled in the art will understand that the principles of the present disclosure may be implemented in any suitably arranged wireless communication systems.

    [0017] FIG. 1 illustrates an example configuration of a full-dimensional multiple input multiple output (FD-MIMO) system 100 according to an exemplary embodiment of the disclosure. The embodiment of the FD-MIMO system 100 shown in FIG.1 is for illustration only. Other embodiments of the FD-MIMO system 100 could be used without departing from the scope of this disclosure.

    [0018] As illustrated in FIG. 1, the FD-MIMO system 100 comprises an FD-MIMO eNB 102, an elevation beamforming 104, an azimuth beamforming 106, and a plurality of user equipments (UEs) 108. Specifically, the FD-MIMO system 100 comprises a 2Dimensional (D) antenna array plane that is deployed with much more antenna elements than traditional multiple antenna systems in a wireless communication system. The antenna elements allow dynamic and adaptive precoding to be performed jointly across all antennas. As a result of such precoding, the FD-MIMO eNB 102 (such as base station) achieves more directional transmissions with the azimuth beamforming 106 and the elevation beamforming 104 simultaneously to the plurality of UEs 108.

    [0019] Depending on the network type, other well-known terms may be used instead of "eNodeB" or "eNB," such as "base station" or "access point." For the sake of convenience, the terms "eNodeB" and "eNB" are used in this patent document to refer to network infrastructure components that provide wireless access to remote terminals. Also, depending on the network type, other well-known terms may be used instead of "user equipment" or "UE," such as "mobile station," "subscriber station," "remote terminal," "wireless terminal," or "user device." For the sake of convenience, the terms "user equipment" and "UE" are used in this patent document to refer to remote wireless equipment that wirelessly accesses an eNB, whether the UE is a mobile device (such as a mobile telephone or smartphone) or is normally considered a stationary device (such as a desktop computer or vending machine).

    [0020] One or more of the components illustrated in FIG. 1 may be implemented in specialized circuitry configured to perform the noted functions, or one or more of the components may be implemented by one or more processors executing instructions to perform the noted functions.

    [0021] FIG. 2 illustrates an example message flow of an FD-MIMO system 200 according to an exemplary embodiment of the disclosure. The embodiment of the FD-MIMO system 200 shown in FIG.2 is for illustration only. Other embodiments of the FD-MIMO system 200 could be used without departing from the scope of this disclosure.

    [0022] As illustrated in FIG. 2, the message flow of the FD-MIMO system 200 comprises an eNB 210, a UE1 220, a UE2 230, a plurality of common reference signal (CRS) and common control signals 212, 214, a sounding reference signal (SRS) 222 from the UE1 220, an SRS 232 from the UE2 230, a plurality of physical downlink shared channels (PDSCH) 216, 218, an acknowledge/negative acknowledge (ACK/NACK) signal 224 from the UE1 220, and an ACK/NACK 234 signal from the UE2 230. The eNB 210 calibrates an antenna and a transceiver, and then calculate an antenna virtualization. The eNB 210 sends the plurality of CRS and common control signal 212, 214 including the calculation results of the antenna virtualization to the UE1 220, UE2 230. The UE1 220 and the UE2 230 configure a SRS and uplink channel for the eNB 210, respectively. The UE1 220 and the UE2 230 send the SRS to the eNB 210, respectively. After receiving the SRSs 222, 232 from the UE1 220 and the UE2 230, the eNB 210 estimates an SRS channel based on the SRS 222, 232 transmitted from the UE1 220 and UE2 230, and the eNB 210 performs a channel quality indication prediction, a modulation and coding scheme (MCS), a scheduling, and a precoding. The eNB 210 sends the plurality of PDSCHs 216, 218 to the UE1 220 and the UE2 230, respectively. The UE1 220 and the UE2 230 receive data transmitted on the PDSCH 216, 218, respectively. Finally, the UE1 220 and the UE2 230 send the ACK/NACK signals 224, 234, respectively, to the eNB 210.

    [0023] The operations of FD-MIMO systems make a provision for achieving a higher data rate and a high-order multi-user MIMO (MU-MIMO) by utilizing a 2D antenna array. In certain embodiments, an FD-MIMO base station is deployed with 2D antenna array comprising of many more antenna elements than traditional multiple antenna systems. In such embodiments, the FD-MIMO system leads to the impressive improvement on system throughputs and supports the higher-order MU-MIMO.

    [0024] In certain embodiments, an antenna array virtualization creates wide beams required for common control signals and broadcasting signals in a wireless communication system. Those common control signals and broadcasting signals include a cell-specific reference signal (CRS), a channel state information reference signal (CSIRS), a physical broadcast channel (PBCH), and a primary and secondary synchronization signals (PSSS and SSS). An amount of output power of those common channels is assured by activating all elements of the 2D antenna array in a 2D antenna array virtualization technique. In such embodiments, there are two aspects highlight the 2D antenna array virtualization in an FD-MIMO system operation. In one embodiment, an 2D antenna array virtualization is performed on any channel as needed that occupies a part or whole time-frequency resource to comprise mixed precoding symbols with other beamformed data channel. In another embodiment, a 2D antenna array virtualization operates in a flexible way by only driving single antenna, or using an amplitude taper scheme, or activating all antenna elements evenly.

    [0025] In certain embodiments, an FD-MIMO system is used to overcome the challenge for a higher-order MU-MIMO and beamforming problems. In one embodiment, a channel quality indication (CQI) prediction is used to bridge a gap caused by different beam forming schemes such as a virtualized wide beam and a dedicated beam for each user. In another embodiment, a demodulation reference signal (DMRS) mapping provides a feasible operation to accommodate a higher-order MU-MIMO within the current standardized framework (such as 3GPP LTE). In yet another embodiment, per RB based precoder generation is used to counter a frequency selective channel reality in a wide band wireless communication system. The per RB based precoder generation balances a processing complexity and necessity to avoid a degeneration caused by a multipath fading effect.

    [0026] In certain embodiments, a hardware calibration such as a full and partial (or separate) transmission and reception chains measurement is used to enhance performance of an FD-MIMO. Using calibration information, channel state information (CSI) in the air for each UE is precisely estimated, and, by using a reciprocity property of a TDD channel, beamforming precoders are applied to each UE's data traffic as well as a DMRS channel. The calibration information is also used to provide essential elements for antenna port virtualization precoders.

    [0027] One or more of the components illustrated in FIG. 2 may be implemented in specialized circuitry configured to perform the noted functions, or one or more of the components may be implemented by one or more processors executing instructions to perform the noted functions.

    [0028] FIG. 3 illustrates an example hardware (HW) configuration 300 of an FD-MIMO system according to an exemplary embodiment of the disclosure. The embodiment of the HW configuration 300 shown in FIG.3 is for illustration only. Other embodiments of the HW configuration 300 could be used without departing from the scope of this disclosure.

    [0029] As illustrated in FIG. 3, the HW configuration 300 comprise a 2D antenna array 302 (such as 32 RF front-end), a dedicated calibration circuit 304, a baseband signal processing unit 306, and a plurality of common public radio interface (CPRI) 308. The baseband signal processing unit 306 includes RF units, a baseband analog circuit connected with a baseband processing unit through the CPRI 308.

    [0030] As illustrated in FIG. 3 the 32 RF front-ends 302 are distributed on 4 physically independent boards that are connected to the baseband signal processing unit 306 through the 4 CPRI connections 308. The 4 CPRI connections introduce a sample level mismatching. Accordingly, a coarse alignment is designed to detect and compensate the sample level mismatching across multiple CRPI connections.

    [0031] One or more of the components illustrated in FIG. 3 may be implemented in specialized circuitry configured to perform the noted functions, or one or more of the components may be implemented by one or more processors executing instructions to perform the noted functions.

    [0032] FIG. 4 illustrates an example unit diagram of an baseband processing 400 for an FD-MIMO system according to an exemplary embodiment of the disclosure. The embodiment of the base band processing system 400 shown in FIG.4 is for illustration only. Other embodiments of the base band processing system 400 could be used without departing from the scope of this disclosure.

    [0033] As illustrated in FIG. 4, the base band processing system 400 comprises a SRS based channel estimation (per RB) unit 402, a precoder update unit 404, a scrambling and modulation unit 406, an uplink processing unit 408 including a CQI processing, a virtualization unit 410 (such as CRS, CSRS, PSS, SSS), a physical downlink shared channel (PDSCH) precoding unit 412, a DMRS processing and mapping unit 414, a combining unit 416, a synchronization, a cyclic prefix (CP) removing and fast fourier transform (FFT) unit 418, a CP insertion and inverse FFT(IFFT) unit 420, an analog to digital converting (ADC) units 422, a digital to analog converting (DAC) unit 424, and an RF front-end unit 426. An estimated signal of the SRS based channel estimation unit 402 is delivered to the virtualization unit 410, the precoding unit 412, and the DMRS processing and mapping unit 414 through the precoder update unit 404. The combining unit 416 combines output signal from the virtualization unit 410, the precoding unit 412, and the DMRS processing and mapping unit 414. The combined signal of the combining unit 416 is delivered to the DACs unit 424 though the CP insertion and IFFT unit 420, and then transmitted to the RF front-end unit 426 to be transmitted to a receiver though the DACs unit 424. For a receiving operation, the RF front-end unit 426 delivers signals received from the transmitter to the synchronization, CP removing and FFT unit 418 through the ADCs unit 422. And then the output of the synchronization, CP removing and FFT unit 418 delivered to the SRS based channel estimation unit 402 through the uplink processing unit 408 including a CQI prediction.

    [0034] One or more of the components illustrated in FIG. 4 may be implemented in specialized circuitry configured to perform the noted functions, or one or more of the components may be implemented by one or more processors executing instructions to perform the noted functions.

    [0035] FIG. 5 illustrates an example flowchart of a basic processing method 500 for an FD-MIMO system according to an exemplary embodiment of the disclosure. The embodiment of the FD-MIMO basic processing method 500 shown in FIG.5 is for illustration only. Other embodiments of the basic processing method 500 for the FD-MIMO system could be used without departing from the scope of this disclosure.

    [0036] As illustrated in FIG. 5, the basic processing method 500 begins at step 502. Subsequently, the method 500 proceeds to step 504, where a controller detects an instance of a radio frame. If the controller detects the radio frame, the method 500 proceeds to step 506. If not, the method proceeds to step 504. Subsequently, the method 500 proceeds to step 506, where the controller performs a calibration operation. If the calibration operation is skipped, the method 500 proceeds to step 512. If not, the controller proceeds to step 508, where the controller performs a transmit (Tx) calibration if the controller processes to transmit signals to the receiver. In contrast, the method 500 proceeds to step 510, where the controller performs receive (Rx) calibration if the controller processes to receive signals from the receiver. Subsequently, the method 500 proceeds to step 512, where the controller performs a synchronization, a CP removing, and an FFT processing for the signal. Subsequently, the method 500 proceeds to step 514, where the controller performs a physical uplink shared channel (PUSCH) and a physical uplink control channel (PUCCH) symbol processing if the controller receives signals from the receiver. Subsequently, the method 500 proceeds to step 516, where the controller performs an SRS based uplink channel estimation. Subsequently, the method 500 proceeds to step 518, where the controller performs updating a precoder based on the estimated information.

    [0037] Subsequently, the method 500 proceeds to step 520, where the controller performs a scrambling, a modulation, and a frequency domain symbol generation. Subsequently, the method 500 proceeds to step 522, where the controller performs a precoding for a PDCCH and a PDSCH if the controller transmits signals to the receiver. Next, the method 500 proceeds to step 524, wherein the controller performs virtualization for a CRS, a CSRS, a PSS, and an SSS if the controller transmits signals to the receiver. Thereafter, the method 500 proceeds to step 526, where the controller performs a DMRS processing. Finally, the method 500 proceeds to step 528, where the controller performs an IFFT and a CP insertion.

    [0038] One or more of the components illustrated in FIG. 5 may be implemented in specialized circuitry configured to perform the noted functions, or one or more of the components may be implemented by one or more processors executing instructions to perform the noted functions.

    [0039] Table 1 and Table 2 show simulation assumptions and configurations. As shown in Table 1 and Table 2, different antenna array architectures with a baseline LTE eNB antenna array configuration is simulated.


    Simulation Setup:



    [0040] 
    • 3D ITU, UMa
    • 57 sectors with K=10/25 UEs per sector
    • Center frequency 2GHz, bandwidth 10MHz
    • UE speed 3km/h
    • 20% outdoor, 80% indoor UEs
    • UE: 2 Rx(H-V-pol)
    • 8S: X-pol, down-tilt 12-deg




    [0041] Table 1 and Table 2 show two different 4 FD-MIMO antenna array architectures, for example, two different antenna array configurations such as 0.5λ and 2λ antenna element spacing in elevation, respectively. As shown in Table 2, specific antenna parameters in conjunction with a 3D spatial channel model (SCM) obtain an average cell throughput gain of approximately 4 times and 8.2 times cell edge throughput gain compared with the LTE system.

    [0042] FIG. 6 illustrates an example configuration of a 4 FD-MIMO antenna array architecture 600 for a simulation according to an exemplary embodiment of the disclosure. The embodiment of the FD-MIMO antenna array architecture 600 shown in FIG.6 is for illustration only. Other embodiments of the FD-MIMO antenna array architecture 600 could be used without departing from the scope of this disclosure.

    [0043] As illustrated in FIG. 6, the 4 FD-MIMO antennas array architecture 600 for antenna polarization arrangements comprise a cross-polarized array 602 referred to as X-pol and an alternating polarized array 604 referred to as Alt-pol. A result of system level simulation shown in Table 2 is obtained using those two antennas array configurations illustrated in FIG. 6.

    [0044] One or more of the components illustrated in FIG. 6 may be implemented in specialized circuitry configured to perform the noted functions, or one or more of the components may be implemented by one or more processors executing instructions to perform the noted functions.

    [0045] FIG. 7 illustrates an example configuration of an FD-MIMO 2 dimensional (D) antenna array 700 according to an exemplary embodiment of the disclosure. The embodiment of the FD-MIMO 2D antenna array 700 shown in FIG.7 is for illustration only. Other embodiments of the FD-MIMO 2D antenna array 700 could be used without departing from the scope of this disclosure.

    [0046] As illustrated in FIG. 7, the large scale FD-MIMO 2D antenna array 700 comprises 4 vertically arranged panels 702, 704, 706, 708, a sub-array 710, and a patch elements 712, 714, 716, 718. Each of the vertically arranged panels 702, 704, 706, 708 includes eight sub-arrays (such as sub-array 710) each of which is arranged in an 8H x 1V configuration. Each of the sub-array 710 comprises the patch elements 712, 714, 716, 718 fed with a corporate feed network in a 1 horizontal (H) x 4 vertical (V) configuration.

    [0047] In certain embodiments, an FD-MIMO array includes ±45° rotated patch antenna elements that yield dual-linear polarization on two diagonal planes (such as ϕ=±45° as illustrated in FIG. 7). In such embodiments, the +45° and -45° sub-arrays have the same beam widths in an elevation (ϕ=0°) domain and an azimuthal (ϕ=90°) domain. Specifically, the +45° and -45° sub-arrays are interlaced (such as orthogonally polarized) across each of vertically arranged panel (such as 4 vertically arranged panels 702, 704, 706, 708) in both array dimensions (such as H and V) to increase an isolation between adjacent sub-arrays (such as 710).

    [0048] In certain embodiments, the patch elements 712, 714, 716, 718 of the sub-array 710 are fed through a corporate microstrip line feed network designed at a bottom layer of a ground plane. Therefore, energy is coupled to the each of patch elements 712, 714, 716, 718 through rectangular slot openings on the ground plane. In such embodiments, a feeding technique is provided for better bandwidth and higher isolation between the adjacent patch elements 712, 714, 716, 718, as compared to a direct probe feeding. An air gap between an antenna board and the ground plane is selected to maximize a bandwidth and a gain.

    [0049] In such embodiments, performance of the measured sub-array is obtained with a polarization (such as dual-linear ±45°), a bandwidth (such as 2.496-2.69GHz), a beam width (such as 24°-64° for an elevation and azimuth), a gain (such as 10dBi), and a return loss (such as >12dB).

    [0050] One or more of the components illustrated in FIG. 7 may be implemented in specialized circuitry configured to perform the noted functions, or one or more of the components may be implemented by one or more processors executing instructions to perform the noted functions.

    [0051] FIG. 8 illustrates an example unit diagram of an FD-MIMO antenna array visualization 800 according to an exemplary embodiment of the disclosure. The embodiment of the FD-MIMO antenna array virtualization 800 shown in FIG.8 is for illustration only. Other embodiments of the FD-MIMO antenna array virtualization 800 could be used without departing from the scope of this disclosure.

    [0052] As illustrated in FIG. 8, the FD-MIMO antenna array virtualization 800 comprises a requirement unit 802, 2D antenna array virtualization precoder generation units 804, 806, 808, 810, and 2D virtualization units 812, 814, 816, 818. The requirement unit 802 provides requirements to input channels (such as dedicated channels, CRS, CSIRS, PSS, SSS, and PBCH) each of which requires different requirements for virtualizations. Based on the requirements provided by the requirement unit 802, each of channels is processed by the antenna array virtualization precoder generation units 804, 806, 808, 810. For example, the CRS is processed by the CRS dedicated 2D antenna array virtualization precoder generation unit 804. Furthermore, each of channels that have been processed by the dedicated 2D antenna array virtualization precoder generation units 804, 806, 808, 810 is processed by the dedicated 2D virtualization units 812, 814, 816, 818. As similar with the 2D antenna array virtualization precoder generation units 804, 806, 808, 810, the 2D virtualization units 812, 814, 816, 818 are dedicated into the each of channels. For example, the CRS that has been processed by the 2D antenna array virtualization precoder generation unit 804 is processed by the 2D virtualization unit 812.

    [0053] In certain embodiments, an antenna virtualization scheme integrates channels with different beam widths and patterns. In certain embodiments, virtualized channels are combined into symbols in order to generate a mixed beamforming pattern and/or an overlapped beamforming pattern in time domain. As aforementioned, an antenna virtualization in a wireless communications system is used to generate a radiation beam with an expected beam width and a pattern by transmitting a precoded data stream to an antenna array. In addition, the antenna virtualization requires 3-D beams. In certain embodiments, a virtualized beam is activated by only single antenna, a part of antennas following some amplitude taper schemes, or all the antenna elements to provide power control gain at a system level.

    [0054] One or more of the components illustrated in FIG. 8 may be implemented in specialized circuitry configured to perform the noted functions, or one or more of the components may be implemented by one or more processors executing instructions to perform the noted functions.

    [0055] FIG. 9 illustrates an example configuration of an FD-MIMO antenna array visualization 900 according to an exemplary embodiment of the disclosure. The embodiment of the FD-MIMO antenna array visualization 900 shown in FIG.9 is for illustration only. Other embodiments of the FD-MIMO antenna array visualization 900 could be used without departing from the scope of this disclosure.

    [0056] As illustrated in FIG. 9, the FD-MIMO antenna array virtualization 900 comprises a time domain 902, a frequency domain 904, a plurality of resource units 906, a CRS beam 908, a CSIRS beam 910, a PSS beam 912, a SSS beam 914, a PBCH beam 916, a UE specific reference signal (UERS) beam 918, and a PDSCH beam 920. As aforementioned, each of the channels (such as CRC, CSIRS, PSS, SSS, PBCH, UERS, and PDSCH) is transmitted on each of resource units 904 that is determined by the time domain 902 (such as time resources) and the frequency domain 904 (such as frequency resources). In addition, each of channels is processed by each of the antenna array virtualization precoder generation units 804, 806, 808, 810 and each of the virtualization units 812, 814, 816, 818, respectively. Each of the channels (such as CRC, CSIRS, PSS, SSS, PBCH, UERS, and PDSCH) is transmitted with each of beam patterns.

    [0057] One or more of the components illustrated in FIG. 9 may be implemented in specialized circuitry configured to perform the noted functions, or one or more of the components may be implemented by one or more processors executing instructions to perform the noted functions.

    [0058] FIG. 10 illustrates an example configuration of an FD-MIMO beam forming in a 2D large scale antenna array 1000 according to an exemplary embodiment of the disclosure. The embodiment of FD-MIMO beam forming in a 2D large scale antenna array 1000 shown in FIG.10 is for illustration only. Other embodiments of the FD-MIMO beam forming in a 2D large scale antenna array 1000 could be used without departing from the scope of this disclosure.

    [0059] As illustrated in FIG. 10, the FD-MIMO beam forming in a 2D large scale antenna array 1000 comprises an excite only one element beam forming 1005, an excite few elements beam forming with an amplitude taper 1010, and an excite all elements beam forming with a phase taper 1015. Specifically, the excite few elements beam forming with an amplitude taper 1010 includes an aggressive amplitude taper and the excite all elements beam forming with a phase taper 1015 includes a very right amplitude taper and different phases.

    [0060] In certain embodiments, a beam is transmitted from one antenna array elements to generate a wide beam that covers a specific sector angle using a 2D active antenna array. However, the beam is transmitted with very little power because only one element (such as out of typically a few decades of elements) is excited. Thus a beam range is limited and a maximum power rating for transmission per element is consumed.

    [0061] In certain embodiments, a beam is transmitted from a few antenna elements (such as 1005 and 1010) each of which includes a certain amplitude weight (such as pattern synthesis). As noted, FIG. 10 shows a 1-D beam covering a planar sector (such as azimuth), and the beam is visualized as 2D beam covering a sector with specific elevation and azimuth dimensions. In such embodiments, a pattern synthesis scheme for control of side lobes and control of precise beam widths result in an amplitude tapers that heavily excite a few antenna elements at the center of the antenna array while the majority of the elements is left unexcited. Accordingly, power is lost and a beam range is limited while consuming a maximum power rating at each antenna element.

    [0062] In certain embodiments, large scale 2D FD-MIMO antenna arrays provide many active antenna elements and reduce a maximum power rating per element, but still needs to maintain a higher total transmitted power. In such embodiments, generating a 2D beam with specific beam widths in an elevation and azimuth is not trivial. Furthermore, when only a few elements within a large antenna array are excited to create a specific side beam, the wear and tear these elements experience as compared to the rest of the antenna elements is significant, leading to reliability issues and potential hardware failure in the long run.

    [0063] In certain embodiments, all antenna elements are uniformly excited in an amplitude (or at least with a very small taper) with a different phase profile (such as phase taper) so as to create a wide beam and control side lobe levels. In such embodiments, generating a wide beam with all antenna elements excited at full power is not a trivial task since, in general, a fully excited antenna array generates a focused narrow beam width pattern (as illustrated in FIG. 11 and FIG. 3).

    [0064] One or more of the components illustrated in FIG. 10 may be implemented in specialized circuitry configured to perform the noted functions, or one or more of the components may be implemented by one or more processors executing instructions to perform the noted functions.

    [0065] FIG. 11 illustrates an example performance result of an FD-MIMO 2D antenna array 1100 according to an exemplary embodiment of the disclosure. The embodiment of the performance results of an FD-MIMO 2D antenna array 1100 shown in FIG.11 is for illustration only. Other embodiments of the simulation results of an FD-MIMO 2D antenna array 1100 could be used without departing from the scope of this disclosure.

    [0066] As illustrated in FIG. 11, the performance results 1100 comprises performance results of a 2D FD-MIMO antenna array with a beamformed gain pattern 1105, performance results of an amplitude taper based pattern synthesis 1100, and performance results of a wide beam pattern using uniform array excitation and phase taper 1115 (such as all antenna array elements excited with equal amplitudes but different phases). Specifically, the performance results of an amplitude taper based pattern synthesis 1100 synthesizes a wide beam pattern with specific beam widths in elevation and azimuth using an aggressive amplitude taper, where the majority of antenna array elements are almost un-excited. In certain embodiments, the FD-MIMO illustrated in FIG. 3 is implemented with an amplitude taper scheme to generate a wide beam pattern illustrated in Figure 11 (such as 1110).

    [0067] As illustrated in FIG. 11, the pattern synthesis 1100 based on the amplitude taper results in heavy power loss compared to a uniformly excited array (such as 1115). Accordingly, in order to generate a wide beam with certain beam widths in azimuth and elevation and all antenna elements that are equally excited (or almost equally), a certain phase taper is applied. In addition, the certain phase taper has a robustness to account for possible phase errors in a phase calibration process.

    [0068] As aforementioned, because a desired 2D beam pattern is generated with a total transmitted power that is equal to a maximum transmitted power, a beam range is maximized. As illustrated in FIG. 11 (such as 1115), a 3dB beam width covers a sector of about 65deg. in azimuth and 10deg. in elevation.

    [0069] One or more of the components illustrated in FIG. 11 may be implemented in specialized circuitry configured to perform the noted functions, or one or more of the components may be implemented by one or more processors executing instructions to perform the noted functions.

    [0070] FIG. 12 illustrates an example unit diagram of an eNodeB (eNB) processing chain with a multi-user channel quality indication (MU-CQI) prediction 1200 according to an exemplary embodiment of the disclosure. The embodiment of eNB processing chain with the MU-CQI prediction 1200 shown in FIG.12 is for illustration only. Other embodiments of eNB processing chain with the MU-CQI prediction 1200 could be used without departing from the scope of this disclosure.

    [0071] As illustrated in FIG. 12, the eNB processing chain with the MU-CQI prediction 1200 comprises an MU-MIMO scheduling and adaptive modulation coding (AMC) unit 1202, an MU-MIMO precoding unit 1204, a channel status information-reference signal (CSI-RS) mapping unit 1206, a 2D transceiver array unit 1208, a feedback and SRS processing unit 1210, and a plurality of antennas 1212. Output signals from the MU-MIMO scheduling and AMC unit 1202 are delivered to the MU-MIMO precoding unit 1204, where a number of signals to be transmitted to the MU-MIMO precoding unit 1204 are determined by a number of MU-MIMO UEs communicating with the eNB. For example, 8 MU-MIMO UEs are being served in a wireless communication network, a total of 8 output signals from the MU-MIMO scheduling and AMC unit 1202 are transmitted to the MU-MIMO precoding unit 1204. Signals from the MU-MIMO precoding unit 1204 are transmitted to the 2D transceiver array unit 1208 through a plurality of transmit resource units (TxRUs). In this example, a total of 32 TxRUs are connected to the 2D transceiver array unit 1208.

    [0072] The CSI-RS mapping unit 1206 transmits input signals through CSI-RS ports to the 2D transceiver array unit 1208, where a number of CSI-RS ports are determined by a number of CSI-RS. In this example, a total of 12 CSI-RS ports are determined. In addition, the feedback and SRS processing unit 1210 transmits signals to the 2D transceiver array unit 1208 while providing feedback information to the MU-MIMO scheduling and AMC unit 1202. Specifically, the feedback and SRS processing unit 1210 performs an MU-CQI predication. Finally, the 2D transceiver array unit 1208 combine and process all of signals from the CSI-RS mapping unit 1206, the MU-MIMO precoding unit 1204, and the feedback and SRS processing unit 1210, and then transmit signals to the plurality of MU-MIMO UEs through the plurality of antennas 1212. In this example, a total of 128 antenna elements are determined.

    [0073] In certain embodiments, a special precoder (such as antenna virtualization) is used for control symbols to ensure a wide coverage. For example, w0=[w1,...,wNt] is defined as the antenna virtualization precoder, a control symbol

    is determined in accordance with equation (1)



    [0074] In an FD-MIMO, an antenna array is 2Dimensional and has many active antenna elements. Therefore, it is non-trivial to design wo to maintain a similar wide-beam pattern as a conventional MIMO system.

    [0075] A CQI is a feedback parameter from UEs that informs an eNB an overall signal-to-noise ratio (SNR) at the UEs, and considerably impacts a transmission scheme, a modulation and coding scheme selected by the eNB. In one example of LTE/LTE-A, a UE usually derives a CQI based on symbols transmitted by an antenna virtualization. In contrast, data symbols are usually precoded by beams with a narrow width to reduce interference for unintended UEs. Therefore, CQI does not match with the SNR of the data channel due to the precoding difference. In an FD-MIMO system, such mismatch is significant since a precoding for data symbols has much narrower beam widths due to a large number of antennas. Therefore, the eNB needs to estimate SNR for data channels based on the feedback CQI (such as CQI prediction).

    [0076] One or more of the components illustrated in FIG. 12 may be implemented in specialized circuitry configured to perform the noted functions, or one or more of the components may be implemented by one or more processors executing instructions to perform the noted functions.

    [0077] FIG. 13 illustrates an example unit diagram of a multi-user channel quality indication (MU-CQI) prediction 1300 according to an exemplary embodiment of the disclosure. The embodiment of the MU-CQI prediction 1300 shown in FIG.13 is for illustration only. Other embodiments of the MU-CQI prediction 1300 could be used without departing from the scope of this disclosure.

    [0078] As illustrated in FIG. 13, the MU-CQI prediction 1300 comprises a feedback report unit 1305, an SRS channel estimation unit 1310, a single user CQI (such as signal to interference noise ratio (SINR)) compensation unit 1315, an MU-CQI (SINR) compensation unit 1320, a precoding and scheduling unit 1325, and an adjusted CQI and MCS unit 1330. A feedback single (such as CQI, rank, precoding matrix index (PMI)) and an SRS channel estimated signal are delivered from the feedback report unit 1305 and the SRS channel estimation unit 1310, respectively, to the SU CQI compensation unit 1315. The SU CQI compensation unit 1315 accounts for a difference between the SINR for SU CQI, and the estimated SRS channel and the feedback report, and then compensates the SU CQI. Similarly, an output signal of the SU CQI compensation unit 1315 and an output signal of the precoding and scheduling unit 1325 are accounted for the MU CQI at the MU-CQI compensation unit 1320. And then, an output signal (such as compensated MU-CQI signal) of the MU-CQI compensation unit 1320 is adjusted to an actual CQI with an MCS level at the adjusted CQI and MCS unit 1330.

    [0079] In certain embodiments, an SINR (or Tx CQI) prediction is performed for an SU-MIMO UE with a 1-Tx antenna. In such embodiments, a predicting SINR scheme assumes no intra-cell interference. An eNB knows virtualization weights applied in downlink common control channels as well as each of channels of individual antennas based on an uplink SRS measurement. Therefore, the eNB reconstructs the downlink channels for common control channels (such as cell-specific reference signal (CRS)) where antenna virtualization is applied. In addition, the eNB accounts for a difference between an SINR of CRS and an actual data channel, and compensates a CQI. Received downlink signals (yk) with the antenna virtualization (wo) at a UE k (assuming such as a single Tx antenna UE) is represented in accordance with equation (2):


    where hk is the channel direction vector for the UE k, that is estimated at the eNB utilizing SRS transmitted by the UE's 1-Tx antenna, so is a transmission symbol, and nk is noise at the UE receiver.

    [0080] For simplicity, it is assumed that feedback CQI,ρ0k, fed back by the UE k is equal to the corresponding SINR estimated at the UE in accordance with equation (3):


    where

    is the receiver noise variance that is unknown to the eNB.

    [0081] When a UE-specific precoder wk is applied, the downlink Tx SINR ρk for data symbols is calculated in accordance with equation (4):



    [0082] As the eNB is aware of the channel direction vector hk via SRS channel estimates, the SINR is obtained for data channels (or Tx CQI) in accordance with equation (5):


    where ρ0k is a feedback CQI, wo is an antenna virtualization precoder, and wk is the UE-specific precoder wk.

    [0083] Once Tx CQI is obtained, the Tx CQI is used for a link adaptation (such as for determining MCS for the UE). Figure 14 shows the simulation results of the prediction scheme achieving 10% normalized prediction error according to an exemplary embodiment of the disclosure.

    [0084] In certain embodiments, a Tx-CQI prediction for an MU-MIMO is performed for a 1-antenna UE with a 1 CQI and a 1 SRS. When MU-MIMO operation exists, an eNB needs to not only compensate for an SINR mismatching between a CRS (common control signal) and a UE-RS (data signal) but also accounts for an MU interference. In such embodiments, 3 types of MU SINR compensation schemes are considered as shown in Table 3.
    Table 3:
    AlternativesComplexityPerformanceRemark
    Power-reduced SU SINR Little power normalization needed Low No MU interference captured, over-estimated SINR with large mismatch
    MRC MU SINR Medium calc. inner product Medium MU interference captured; Under-estimated SINR as Rx is MMSE IRC
    MMSE MU SINR High matrix inversion needed High MU interference captured; Well matched SINR


    [0085] In the power-reduced SU SINR scheme as shown in Table 3, the predicted SU SINR is divided by a number of MU-MIMO UEs that are co-scheduled. This scheme is simple and accounts for the fact that the power is equally shared among different MU-MIMO UEs. Accordingly, the SINR is reduced proportionally.

    [0086] In the maximum-ratio combining receiver (MRC) MU SINR scheme as when in Table 3, an eNB assumes that the MRC is used at UEs. Because the eNB knows a data precoding and channels for different UEs, and therefore, the eNB estimates MU interference. More specifically, the eNB follows three steps to calculate the MU SINR. At step 1, the eNB maps CQI into an SINR po based on a certain mapping rule. At step 2, the eNB obtains channel estimates based on one SRS, denoted by µlhl, where hl (for 1 transmit antenna and Nr receive antennas) is the 1×Nr normalized channel direction vector and µl is the power associated with this channel for UE 1. Due to a CQI mismatching, a downlink channel SNR (such as power) is different than an uplink SNR estimated with SRS. At step 3, the eNB reconstructs (such as re-calculate) the SINR or an MU-CQI based on an SRS channel estimation. In this example, the MU SINR (such as a number of co-scheduled MU UEs (L) at the Rx of UE l) is predicted in accordance with equation (6):

    where P denotes the total transmitted power at the eNB,

    denotes noise power at the UE 1, and wl is the precoding vector for the l-th UE.

    [0087] It is assumed that the UE computes its SINR (CQI) under a hypothesis that an eNB employs a conjugate beamforming with the total transmitted power P (such as the precoding vector of UE 1 is equal to

    where H denotes Hermitian operation). The SINR computed by the UE is termed as a single user SINR (or feedback CQI) and given by equation (7):



    [0088] For conjugate beamforming, the MU-SINR (such as Tx CQI) is computed in accordance with equation (8):

    where the correlation coefficient ρli is defined as



    [0089] Once L Tx CQI's are obtained for the L MU-MIMO UEs, the Tx CQI is used for a link adaptation (such as for determining MCS for each UE participating in the MU-MIMO transmission). In order to support up to 4 UE MU-MIMO operations, a mapping between a UE and an antenna port is required.

    [0090] In certain embodiments, a scrambling ID (SCID) is combined with a DMRS port to support 4-UE MU-MIMO. An exemplary mapping is given in Table 4.
    Table 4
    UE #Port AssignmentSCID
    UE1 DMRS Port 7 0
    UE2 DMRS Port 8 0
    UE3 DMRS Port 7 1
    UE4 DMRS Port 8 1


    [0091] In order to support up to 8-UE MU-MIMO operations, 8 layers are mapped with 8 UEs each of which includes 1 layer transmission. An exemplary mapping is given in Table 5.
    Table 5:
    UE #Port AssignmentSCID
    UE1 DMRS Port 7 0
    UE2 DMRS Port 8 0
    UE3 DMRS Port 9 0
    UE4 DMRS Port 10 0
    UE5 DMRS Port 11 0
    UE6 DMRS Port 12 0
    UE7 DMRS Port 13 0
    UE8 DMRS Port 14 0


    [0092] In general, the power-reduced SU SINR scheme and the maximum-ratio combining receive MU SINR scheme as shown in Table 3 are combined to increase a number of supportable UEs. For instance, with both SCID=0 and 1, the mapping in Table 5 is extended to support up to 16 UEs. This allows a more efficient DMRS resource allocation and, at the same time, an increased DMRS capacity.

    [0093] In certain embodiments, a precoder is generated with a processing of an output of channel estimation in an FD-MIMO system. Accordingly, a simple conjugate beamforming or more advanced scheme are performed appropriately.

    [0094] FIG. 15 illustrates an example configuration of a sounding reference signal (SRS) channel assignment 1500 for an uplink channel arrangement to support a high-order MU-MIMO in an FD-MIMO system according to an exemplary embodiment of the disclosure. The embodiment of the SRS channel assignment 1500 shown in FIG.15 is for illustration only. Other embodiments of the SRS channel assignment 1500 could be used without departing from the scope of this disclosure.

    [0095] As illustrated in FIG. 15, the SRS channel assignment 1500 comprises a plurality of radio frames 1505, a plurality of downlink sub-frames (D) 1510, a special sub-frame (S) 1515, and an uplink sub-frame (U) 1520. Each of radio frames 1505 includes the plurality of downlink sub-frames 1510, the special sub-frames 1515, and the uplink sub-frames 1520. More specifically, the special sub-frame (S) 1515 and the uplink sub-frame 1520 (U) include a plurality of symbols 1525 (such as four symbols) for an uplink SRS channel.

    [0096] The four symbols 1525 are denoted as SRS 0, SRS 1, SRS 2, and SRS 3. Each symbol is designed to accommodate two SRS channels in the shared channel 1515 and the uplink channel 1520, respectively, for one UE. The two SRS channels are interleaved to form a wideband channel for a channel estimation operation at an eNB side. As illustrated in FIG. 15, a plurality of sub-carriers 1530 that are allocated to the plurality of the SRS channels 1525 are marked with a plurality of arrow end sub-carriers 1535, a plurality of dot end sub-carriers 1540, and a plurality of circle end sub-carriers 1545. The plurality of the arrow end sub-carriers 1535 comprises one SRS channel for one UE. In contrast, the plurality of the dot end sub-carriers 1540 comprises another SRS channel for one UE. The plurality of the null sub-carriers 1545 marked with circle is a direct current (DC) (such as upper and lower side band).

    [0097] As illustrated in FIG. 15, the last two symbols (such as SRS 0 and SRS1) in the special sub-frame 1515 and the last two symbols (such as SRS 2 and SRS 3) in the uplink sub-frame 1520 are designed to allocate an uplink SRS channel. Every other subcarrier are allocated to one UE if the subcarriers are separated by odd and even number subcarriers. Accordingly, the odd and even number subcarriers are assigned to two UEs SRS signal. As a result, a total of eight UEs are supported in the FD-MIMO system operations.

    [0098] One or more of the components illustrated in FIG. 15 may be implemented in specialized circuitry configured to perform the noted functions, or one or more of the components may be implemented by one or more processors executing instructions to perform the noted functions.

    [0099] FIG. 16 illustrates an example flowchart of an SRS based channel estimation and per resource unit (RB) precoder generation method 1600 according to an exemplary embodiment of the disclosure. The embodiment of the SRS based channel estimation and per RB precoder generation method 1600 in FIG. 16 is for illustration only. Other embodiments of the SRS based channel estimation and per RB precoder generation method 1600 could be used without departing from the scope of this disclosure.

    [0100] As illustrated in FIG. 16, the method 1600 begins at step 1602, where the method 1600 estimates a channel per RB. Next the method 1600 proceeds to step 1604, where the method performs fitting operation for the per RB estimated channel. Therefore, per RB estimated channel fitting presents channel information in every RB to trade off implementation complexity and processing accuracy. Subsequently, the method proceeds to step 1606, where the method normalizes the per RB estimated channel. Finally, the method proceeds to step 1608, where the method generates per RB precoder. As illustrated in FIG. 16, the method 1600 performs uplink channel estimation and precoder generation on each sub-carrier based on SRS.

    [0101] In certain embodiments, the normalization at step 1606 illustrated in FIG. 16 is performed on a channel status vector on each RB. Accordingly, the normalization based on the channel status vector of the kth UE on the lth RB as Wkl is obtained in accordance with equation (9)



    [0102] One or more of the components illustrated in FIG. 16 may be implemented in specialized circuitry configured to perform the noted functions, or one or more of the components may be implemented by one or more processors executing instructions to perform the noted functions.

    [0103] FIG. 17 illustrates an example flowchart of a calibration method 1700 for a Tx and an Rx according to an exemplary embodiment of the disclosure. The embodiment of the calibration method 1700 shown in FIG.17 is for illustration only. Other embodiments of the calibration method 1700 could be used without departing from the scope of this disclosure.

    [0104] As illustrated in FIG. 3 (such as system level architecture), 32 RF front-ends are distributed on 4 physically independent boards each of which is connected to the baseband signal processing unit 306 through the 4 CPRI connections 308. Each RF board (such as 32 RF boards) is equipped with a common Tx and Rx channel. Accordingly, a calibration function is achieved across the eights Tx and Rx channels in a single board.

    [0105] In certain embodiments, an auxiliary switch network is designed on each RF board to calibrate the channels on multiple RF boards. The auxiliary switch network is firstly connected to the common channels that transmits and receives signals. In addition, the auxiliary switch network accomplishes a calibration function for all 32 channels (such as 32 RFUs).

    [0106] As illustrated in FIG. 17, the method begins at step 1702. Subsequently, the method 1700 proceeds to step 1704, where the method performs a coarse time alignment to detect and compensate a sample level mismatching across multiple CRPI connections (such as 308 illustrated in FIG.3). As illustrated in FIG. 3 the 32 RF front-end 302 are distributed on 4 physically independent boards that are connected to the baseband signal processing unit 306 through the 4 CPRI connections 308. The 4 CPRI connections introduce the sample level mismatching. Next, the method 1700 proceeds to step 1706, where the method performs a fine time alignment to detect and compensate a delay differences across multiple channels to less than one nanosecond. Subsequently, the method 1700 proceeds to step 1708, where the method 1700 performs a phase alignment to tune a phase of multiple channels into an in-phase. Finally, the method 1700 ends at step 1710.

    [0107] In certain embodiments, an RF front-end calibration measures a gain, a timing, and a phase difference across multiple Rx channels as well as multiple Tx channels. In such embodiments, a precoder generation (such as 1608 as illustrated in FIG. 16) needs the calibration information to compensate an impact of an RF chain. Moreover, the RF front-end calibrations make the antenna port virtualization feasible in an FD-MIMO system.

    [0108] In certain embodiments, filters in an RF front-end introduce a group delay from a few nanoseconds to more than ten nanoseconds. Because an FD-MIMO system is a broad band wireless communications with at least 10 MHz band, a few nanoseconds group delay is not negligible and compensated by only one phase. The fine time alignment 1706 detects and compensates the delay differences across multiple channels to less than one nanosecond.

    [0109] One or more of the components illustrated in FIG. 17 may be implemented in specialized circuitry configured to perform the noted functions, or one or more of the components may be implemented by one or more processors executing instructions to perform the noted functions.

    [0110] FIG. 18 illustrates an example unit diagram of a calibration circuit 1800 for a Tx and an Rx according to an exemplary embodiment of the disclosure. The embodiment of the calibration circuit 1800 shown in FIG.18 is for illustration only. Other embodiments of the calibration circuit 1800 could be used without departing from the scope of this disclosure.

    [0111] As illustrated in FIG. 18, the calibration circuit 1800 comprises a plurality of RF front-ends 1802, a board 1 1810, and a board 2 1820. As illustrated in FIG.3, 32 RF front-ends (such as 1802 illustrated in FIG. 18) are distributed on the 4 physically independent boards (such as the board 1 1820 and the board 2 1820) each of which include 8 RF front-ends (such as 1802 illustrated in FIG. 18). The four boards (such as 1810, 1820 illustrated in FIG. 18) are connected to the baseband signal processing unit 306 through the four CPRI connections 308. The RF front-end 1802 includes a common Tx and a common Rx channel. A calibration function is achieved across the 8 Tx and Rx channels in a single board (such as board 1 and board 2 illustrated in FIG. 18). An auxiliary switch network 1804 is equipped on each of boards 1810, 1820. With the assistance of the auxiliary switch network 1804, the calibration functionality is performed for all 32 channels (such as 32 RF front-ends, RFUs).

    [0112] A large-scale antenna system (such as MIMO, or FD-MIMO) in a wireless communication system refers to a communication system with a large number of transmit antennas (Txs) at BS (such as tens or hundreds of Txs). With a large number of Txs employed at BS, a communication system offers rich spatial degrees of freedom and thus is capable of supporting high-order MU-MIMO transmissions.

    [0113] A precoding scheme is commonly used to suppress intra-user interference in MU-MIMO transmissions and plays a critical role in a system performance. To be more specific, maximum ratio transmission (MRT) (such as conjugate beamforming) has a low implementation complexity and maximizes signal strength of an intended UE. However, the MRT scheme does not take intra-user interference into account. Thus, the MRT does not perform well in the interference-limited scenario, where noise is much weaker than interference. A zero forcing (ZF) precoding scheme attempts to null intra-user interference at the expense of noise enhancement. In the noise-limited scenario where interference is much weaker than noise, the ZF precoding suffers a considerable performance loss.

    [0114] One or more of the components illustrated in FIG. 18 may be implemented in specialized circuitry configured to perform the noted functions, or one or more of the components may be implemented by one or more processors executing instructions to perform the noted functions.

    [0115] FIG. 19 illustrates an example unit diagram of a large-scale antenna system 1900 according to an exemplary embodiment of the disclosure. The embodiment of the large-scale antenna system 1900 shown in FIG.19 is for illustration only. Other embodiments of the large-scale antenna system 1900 could be used without departing from the scope of this disclosure.

    [0116] As illustrated in FIG. 19, the large-scale antenna system 1900 comprises a large-scale transmit antenna system at an eNB 1910, a plurality of channel matrix 1920, and a plurality of receive antenna systems 1930.

    [0117] One or more of the components illustrated in FIG. 19 may be implemented in specialized circuitry configured to perform the noted functions, or one or more of the components may be implemented by one or more processors executing instructions to perform the noted functions.

    [0118] In certain embodiments, N,Mk,K, and Qk are defined as a number of the Tx antennas at eNB, a number of receive antennas (Rxs) at a UE, a number of co-scheduled UEs, and a number of data streams at UE k, respectively, for a large-scale antenna system operation. As illustrated in FIG. 19, at a specific subcarrier used for UE-specific reference signals and data, the received signals at the UE k is obtained in accordance with equation (10):


    where Hk denotes the Mk×N channel matrix between eNB and UE k, Wk denotes the N×Qk precoding matrix for the UE k, xk denotes the transmitted signals at UE k, and nk denotes the additive white Gaussian noise at UE k, i.e., nkCN (0, σ2IMk).

    [0119] In certain embodiments, an SLNR-based precoding technique is used. In such embodiments, a channel matrix is obtained in accordance with equation (11) and (12):





    [0120] A total transmitted power per user is constrained by

    with IQk denoting a Qk×Qk matrix, and the precoding matrix satisfies the two constraints such as

    where Tr(·) denotes the trace of a matrix and

    with Dk denoting a diagonal matrix. The latter constraint is due to the assumption that matched filters are employed at Rxs. In the SLNR-based precoding scheme, the precoding matrix Wk is chosen to maximize the following quantity in accordance with equation (13):


    where ck is defined as Mkσ2/Qk and IN is an N×N identity matrix.

    [0121] Mathematically, finding an optimal precoding matrix Wk is formulated in accordance with equation (14):


    where

    and

    for a diagonal matrix Dk.

    [0122] In particular, when the precoding matrix Wk is a N×1 vector, the SLNRk is re-written in accordance with equation (15):



    [0123] In this case, the optimization problem given in (14) is re-written in accordance with equation (16):



    [0124] The optimal solution wk to the generalized Rayleigh quotient problem is given by wk= the most dominant eigenvector of



    [0125] In certain embodiments, every UE has one Rx antenna (such as M1=M2=...=MK=1). In this case, the channel matrix Hk is a N×1 vector for k=1,...,K. In addition, Q1=Q2=...=QK=1 since 1≤Qk≤Mk for k=1,...,K. It implies c1=c2=...=cK (such as all ck for k=1,...,K are equal to a constant c). Thus, the precoding matrix wk is also a N×1 vector. The precoding vector wk for Mk=1 is given by wk= the most dominant eigenvector of



    [0126] In certain embodiments, every UE receives a single stream (such as Q1=Q2=...=QK=1). The channel matrix is not necessarily a vector. However, the optimal precoding matrix wk is still a N×1 vector, that is given by wk= the most dominant eigenvector of



    [0127] In certain embodiments, there exists an N×N invertible matrix Tk. In such embodiments, the following equations (17) and (18) are satisfied simultaneously:





    [0128] The optimal precoding matrix Wk that maximizes SLNRk in equation 13 given by Wk = ρTk(:, 1: Qk), where ρ is a normalization factor such as



    [0129] The major computational complexity for obtaining the precoding matrix Wk lies in the step for finding the non-singular solution Tk to the classical simultaneous diagonalization problem. Conventionally, the solution to the classical simultaneous diagonalization problem is obtained in accordance with following steps. At step 1, Cholesky factorization is applied for the matrix

    to obtain

    where Lk is a N×N lower triangular matrix. And then compute the matrix

    At step 2, a symmetric QR scheme is applied to compute the Schur decomposition of the matrix

    to obtain

    where Yk is a unitary matrix and ∑k is a diagonal matrix. The non-singular solution Tk to the classical simultaneous diagonalization problem is given by

    Accordingly, the matrix Tk satisfies:

    At the last step, the precoding matrix Wk = ρTk(:,1:Qk).

    [0130] Table 6 shows the steps to obtain a single Cholesky decomposition when all ck are the same, i.e., c1=c2=...=cK.



    [0131] In certain embodiments, a matrix operation (such as Cholesky or Schur decompositions) is implemented by different schemes that have different cons and pros. Table 7 shows assumptions that arithmetic with individual elements has complexity O(1).





    [0132] In Table 7, it is assumed that matrix multiplication and matrix inversion are performed in a straightforward manner. Alternatively, additional computation schemes are not considered to the compute matrix multiplication and matrix inversion. Since all schemes are assumed to adopt the same matrix multiplication and matrix inversion schemes if any, Table 7 is used to investigate the relative difference in terms of computational complexity among different schemes.

    [0133]  In certain embodiments, a UE has one Rx antenna (such as case 1 shown in Table 7). In such embodiments, the precoding vector wk(such as M1=M2=...=MK=1) is obtained in accordance with equation (19):

    where

    with



    [0134] Alternatively, the precoding vector wk is expressed in accordance with equation (20):

    where β is a normalized factor to ensure



    [0135] In such embodiments, a single matrix inversion of a N×N matrix is computed for computing all Kprecoding vectors. The overall computational complexity is O(N3)+O(N2). As shown in Table 7, substantial savings in computational complexity is achieved as compared with the case 1 shown in Table 7 (such as a UE has one Rx antenna). Alternatively, the case 3 (such as generalized NxN invertible matrix) shown in Table 7 is used to solve the case 1 with a computational complexity of O(N3). However, the solution for the case 4 is still much higher than the solution presented in the case 1 for a large value of N, that is due to the fact that the solution in the case 4 involves two times matrix operations of order O(N3).

    [0136] In certain embodiments, a precoding vector wk for the case 1 (such as M1=M2=...=MK=1,) shown in Table 7 is obtained by

    ×the kth column of (cIK+HHH)-1 Alternatively, the precoding vector wk is expressed as wk:=βHH×the kth column of (cIK+HHH)-1 where β is a normalized factor to ensure



    [0137] In such embodiments, only a single matrix inversion of a K×K matrix is needed for computing all Kprecoding vectors. In a large scale antenna system (such as N>>K), the method for the case 2 scheme involves much smaller computational complexity than the method for the case 1 while achieving the identical performance. Therefore, the method for the case 2 is more suitable for a large scale antenna system as compared with the method for the case 1.

    [0138] In certain embodiments, the precoding vector wk (such as the case 2 shown in Table 7, Q1=Q2=...=QK=1) is obtained by wk= the most dominant eigenvector of

    The most dominant eigenvector wk for k=1,...,K is computed in the following steps. At step 1, a symmetric QR scheme is applied to compute the Schur decomposition of the matrix HHH such as HHH=UΛUH, where U is a N×N unitary matrix and Λ is a N×N diagonal matrix with non-negative entries. The QR scheme to compute the Schur decomposition of a N×N symmetric matrix has computational complexity of order O(N3). The matrix (ckIN+HHH)-1 is readily obtained as (ckIN+HHH)-1=UH(CkIN+Λ)-1U. Thus, the computation of (ckIN+HHH)-1 has computational complexity of order O(N3). In the case that all ck are the same (such as equal to c), (cIN+HHH)-1 is only needed to perform once. At step 2,

    is computed that has computational complexity of order O(N2ik Mi). At step 3, the most dominant eigenvector of

    is computed that has complexity of O(N2Mk).

    [0139]  As shown Table 7, the method given for the case 3 reduces the number of matrix inversions by a factor of K in the case that all ck are the same, as compared with the method for the case 3 shown in Table 7.

    [0140] In certain embodiments, a precoding vector wk (such as the case 2 shown in Table 7, Q1=Q2=...=QK=1) is obtained by wk= the most dominant eigenvector of

    where



    [0141] In such embodiments, the most dominant eigenvector wk for k=1,...,K is computed in the following steps. At step 1, a symmetric QR scheme is applied to compute the Schur decomposition of the matrix HHH such as HHH=V∑VH, where V is a M×M unitary matrix and ∑ is a M×M diagonal matrix with non-negative entries, where M=∑kMk. The QR scheme to compute the Schur decomposition of a M×M symmetric matrix has computational complexity of order O(M3). The matrix (ckIM+HHH)-1 is readily obtained as (ckIM + HHH)-1 = VH(ckIN + ∑)-1V. Thus, the computation of (ckIM+HHH)-1 has complexity of order O(M3). In addition, computing HH(ckIM+HHH)-1H requires max(O(NM2),O(N2M)).

    [0142] In the case that all ck are the same (such as equal to c),

    is only needed to be computed once. At step 2, the

    that has computational complexity of order O(N2i≠kMi) is computed. At step 3, the most dominant eigenvector of

    that has complexity of O(N2Mk) is computed.

    [0143] In such embodiments, only a matrix inversion of a M×M matrix is computed. In a large scale antenna system, a number of Txs at a BS, N, is much larger than M. Thus, a method in this embodiment has much smaller computational complexity than one a method for the case 3 shown in Table 7. Similar to the method for the case 3 shown in Table 7, the method given for the case 4 reduces a number of matrix inversions by a factor of K in the case that all ck are the same, as compared with the methods shown in Table 7.

    [0144] In certain embodiments, the precoding matrix Wk (such as the case 3, Generalized NxN invertible matrix) is obtained by using the following steps. At step 1, a thin SVD to HH is applied to obtain the matrices U and ∑ in the thin SVD, where U is a N×M unitary matrix and ∑ is a M×M diagonal matrix with non-negative diagonal entries such that UHU=IN. For M<<N, this step has complexity of O(N2M). At step 2, the following matrix is defined as

    Clearly, the matrix Pk satisfies:

    At step 3, an EVD is applied to the matrix

    to obtain

    where Yk is a unitary matrix and ∑k is a M×M diagonal matrix. The non-singular solution Tk to the classical simultaneous diagonalization problem is given by



    [0145] In certain embodiments, single antenna UEs is extended to multi-antenna UEs using the same principles. In case 3 and case 4 shown in Table 7, an implicit assumption is that all UEs have identical receive SNR

    However, this assumption is not hold in practice as UEs that have different SNRs. In addition, for FDD systems, channel state information (CSI) is obtained via a PMI feedback where only directional information is captured. Assume an eNB knows

    it assumes the signal model for precoding in accordance with equation (21):



    [0146] where hk is normalized channel, and pk and

    are channel amplitude before and after noise normalization, respectively.

    [0147] Accordingly, wk:=αk×the kth column of (H((P)-1+HHH)-1) is obtained. Or equivalently, wk:=αk× the kth column of

    is obtained. Where

    and H is normalized with a unit norm. In one embodiment, H is approximated by PMI and

    is approximated by CQI/SINR feedback from a UE.

    [0148] In certain embodiments, a precoder for the UE k is designed assuming PMI or in general the channel direction for UE k is uk, and the SINR (after mapping of CQI feedback) is ρk. In such embodiments, the precoder is obtained in accordance with wk:=normarlized the kth column of



    [0149] Or equivalently, define

    Accordingly, wk = normarlized the kth column of

    is obtained.

    [0150] In such embodiments, the followings mathematical extensions are applied.
    • Observation 1:

      is a rank one matrix and thus it has only one non-zero eigenvalue.
      Reason: rank

      and

      is a full-rank matrix.
    • Observation 2:

      is the most dominant eigenvector.
      Reason:

      The following equation is obtained:

      where

      Sine hk is a non-zero vector and

      is positive definite, λ is a non-zero eigenvalue and v is the corresponding eigenvector. As

      has only one non-zero eigenvalue,

      is the most dominant eigenvector.
    • Observation 3:

      can be written as

      where


      Reason: Define

      Clearly,


      and

      By matrix inversion lemma, the following equations are obtained:






    [0151] By definition of λ and s and EQ 4,

    Then,

    and

    • Observation 4:

      can be rewritten as


      Reason: From EQ 3, the following equation is obtained:

    • Observation 5: (cIN+HHH)-1HH can be rewritten as HH(cIK+HHH)-1.
      Reason: Notice that

    • Observation 6:

      ×the kth column of (cIK+HHH)-1


    [0152] Reason:

    is just the kth column of (cIN+HHH)-1HH, which is the kth column of HH(cIK+HHH)-1. Notice that the matrix HH(cIK+HHH)-1 only involves an inverse of K×K matrix, which has low computational complexity for a small k. In short, there are four equivalent forms of the SLNR beamforming vectors.









    [0153] In such embodiments, the following mathematical extensions are also applied. Since

    the following equation is obtained:





    [0154] Clearly,

    Hence, SLNRk can be rewritten as

    As SLNRk is a monotonically increasing function of µkk, maximizing SLNRk is equivalent to maximizing

    Thus, the optimization problem given in EQ1 is equivalent to the following problem:



    [0155] At step 1, EVD for the matrix HHH to obtain HHH=XΛXH is applied, where X is an

    unitary matrix and Λ is a diagonal matrix with non-negative diagonal entries. Note that HHH is independent of the UE index. Accordingly, the following equation is obtained as

    where the matrix Pk satisfies:



    [0156] At step 2, EVD to the matrix

    to obtain

    is applied, where Yk is a unitary matrix and ∑k is a diagonal matrix. The non-singular solution Tk to the classical simultaneous diagonalization problem is given by



    [0157] FIG. 20 illustrates operations of a base station according to an exemplary embodiment of the disclosure.

    [0158] Referring to FIG. 20, in a step 2001, the base station receives one or more signals. More specifically, the base station receives the one or more signals including information for beamforming to a plurality of UEs using a FD-MIMO beamforming scheme, wherein the FD-MIMO beamforming scheme includes same time resources and same frequency resources that are co-scheduled to the plurality of UEs.

    [0159] In a step 2003, the base station identifies a time delay of the one or more signals. More specifically, the base station identifies the time delay of the one or more signals associated with one or more antenna arrays that are distributed in the large scale antenna array.

    [0160]  In a step 2005, the base station performs a MU joint beamforming. More specifically, the base station performs the MU joint beamforming on the one or more signals to one or more UEs.

    [0161] FIG. 21 illustrates operations of a UE according to an exemplary embodiment of the disclosure.

    [0162] Referring to FIG. 21, In a step 2101, the UE transmits an uplink signal to the BS. Herein, the uplink signal includes CQI information associated with a reference signal received from the BS.

    [0163] In a step 2103, the UE receives one or more beams from one or more antenna arrays. More specifically, the UE receives the one or more beams from the one or more antenna arrays associated with the BS using a FD-MIMO beamforming scheme.

    [0164] Although the present disclosure has been described with an exemplary embodiment, various changes and modifications may be suggested to one skilled in the art. It is intended that the present disclosure encompass such changes and modifications as fall within the scope of the appended claims.


    Claims

    1. A method for operating a base station, BS, (102) in a wireless communication system, the method comprising:

    Receiving (2001) a plurality of signals including information for beamforming to a plurality of user equipments, UEs, using a full-dimensional multiple-input multiple-output, FD-MIMO, beamforming scheme through at least one time resource and at least one frequency resource that are co-scheduled to the plurality of UEs;

    performing (506, 1700, 2003) a calibration for generating a precoder based on the received plurality of signals, wherein the performing the calibration comprises:

    detecting (1704) a sample level mismatch for a plurality of common public radio interface, CPRI, connections associated with at least one antenna array;

    compensating for (1704) the detected sample level mismatch for the plurality of CPRI connections;

    detecting (1706) delay differences across a plurality of channels after the detected sample level mismatch is compensated;

    compensating for (1706) the detected delay differences across the plurality of channels; and

    adjusting (1708) phases of the plurality of channels to be in-phase after the detected delay differences are compensated; and

    performing (2005) the beamforming on signals to the plurality of UEs based on a precoder obtained from a result of the calibration.


     
    2. The method of Claim 1,
    wherein the calibration is performed on a radio frame to provide a timing alignment across the plurality of common public radio interface, CPRI, connections (308), each of the plurality of CPRI connections for connecting an RF front end (302) of the BS (102) to a baseband unit (306) of the BS (102),
    wherein the timing alignment comprises a coarse timing alignment detecting and compensating the sample level mismatch, and a fine timing alignment achieving a broad band calibration, and wherein the method further comprises:

    performing a virtualization of the at least one antenna array in accordance with a 2 dimensional, 2D, plane associated with a signal generated from the precoder;

    updating the precoder in accordance with an estimation of a plurality of channels, wherein the estimation of the plurality of channels is performed on a per resource block, RB, basis; and

    transmitting the signals at the at least one antenna array to the plurality of UEs using the FD-MIMO beamforming scheme.


     
    3. The method of Claim 2, wherein the the method further comprises:

    obtaining (1315) a single user, SU, channel quality indicator, CQI, based on a feedback report received from a UE among the plurality of UE and an estimated SRS channel;

    obtaining (1320) a multiple user, MU, CQI based on the SU CQI, precoding and scheduling information; and

    identifying (1330) an adjusted CQI and a modulation and coding scheme, MCS, level corresponding to the adjusted CQI based on the MU CQI.


     
    4. The method of Claim 2, wherein the performing the virtualization of the at least one antenna array comprises: performing antenna virtualization precoding using a low complexity precoding scheme that reduces K times of matrix inversion to 1 time and an NxN dimension of matrix inversion to a KxK, wherein the low complexity precoding scheme is determined in accordance with a channel matrix that is approximated by a precoding matrix indicator, PMI, and a signal to interference noise ratio, SINR, that is obtained after mapping of a CQI feedback.
     
    5. A base station, BS (102) in a wireless communication system, the BS comprising:

    at least one transceiver; and

    at least one processor operatively coupled with the at least one transceiver,

    wherein the at least one processor is configured to control to:
    receive (2001) a plurality of signals including information for beamforming to a plurality of user equipments, UEs, using a full-dimensional multiple-input multiple-output, FD-MIMO, beamforming scheme through at least one time resource and at least one frequency resource that are co-scheduled to the plurality of UEs, and

    perform (506, 1700, 2003) a calibration for generating a precoder based on the received plurality of signals,

    wherein, in order to perform the calibration, the at least one processor is configured to control to:

    detect (1704) a sample level mismatch for a plurality of common public radio interface, CPRI, connections associated with at least one antenna array,

    compensate for (1704) the detected sample level mismatch for the plurality of CPRI connections,

    detect (1706) delay differences across a plurality of channels after the detected sample level mismatch is compensated,

    compensate for (1706) the detected delay differences across the plurality of channels, and

    adjust (1708) phases of the plurality of channels to be in-phase after the detected delay differences are compensated, and
    wherein the at least one processor is further configured to control to perform (2005) the beamforming on signals to the plurality of UEs based on a precoder obtained from a result of the calibration.


     
    6. The BS of Claim 5, wherein: the calibration is performed on a radio frame to provide a timing alignment across the plurality of common public radio interface, CPRI, connections (308), each of the plurality of CPRI connections for connecting an RF front end (302) of the BS (102) to a baseband unit (306) of the BS (102),
    wherein the timing alignment comprises a coarse timing alignment detecting and compensating the sample level mismatch, and a fine timing alignment achieving a broad band calibration, and
    wherein the at least one processor is further configured to control to:

    perform a virtualization of the at least one antenna array in accordance with a 2 dimensional, 2D, plane associated with a signal generated from the precoder;

    update the precoder in accordance with an estimation of a plurality of channels, wherein the estimation of the plurality of channels is performed on a per resource block, RB, basis; and

    transmit the signals at the at least one antenna array to the plurality of UEs using the FD-MIMO beamforming scheme.


     
    7. The BS of Claim 6, wherein, the at least one processor is configured to control to:

    obtain (1315) a single user, SU, channel quality indicator, CQI, based on a feedback report received from a UE among the plurality of UE and an estimated SRS channel,

    obtain (1320) a multiple user, MU, CQI based on the SU CQI, precoding and scheduling information; and

    identify (1330) an adjusted CQI and a modulation coding scheme, MCS, level corresponding to the adjusted CQI based on the MU CQI.


     
    8. The method of Claim 1 or the BS of Claim 5, wherein the at least one antenna array of the BS comprises at least one virtualization pattern including at least one time domain symbol and multiple virtualized symbols, and
    wherein the at least one antenna array activates at least one antenna element of the BS.
     
    9. The method of Claim 1 or the BS of Claim 5, wherein a scrambling identification, SCID, is allocated into at least one demodulation reference signal, DMRS, port that is mapped to the plurality of co-scheduled UEs sharing the at least one time resource and the at least one frequency resource.
     
    10. The BS of Claim 6, wherein, in order to perform the virtualization of the at least one antenna array, the at least one processor is configured to control to:
    perform antenna virtualization precoding using a low complexity precoding scheme that reduces K times of matrix inversion to 1 time and an NxN dimension of matrix inversion to a KxK, wherein the low complexity precoding scheme is determined in accordance with a channel matrix that is approximated by a precoding matrix indicator, PMI, and a signal to interference noise ratio, SINR, that is obtained after mapping of a CQI feedback.
     
    11. The method of Claim 4 or The BS of Claim 10, wherein the K is determined as a number of UEs being served and the N is determined as a number of antennas to be used to transmit antenna beams to the UEs.
     
    12. The method of Claim 1 or the BS of Claim 5, wherein the at least one antenna array of the BS comprises a plurality of vertically arranged panels each of which includes a plurality of sub-arrays that is arranged in an n number horizontal x 1 vertical configuration, the each of sub-arrays including a plurality of patch elements fed with a corporate feed network.
     


    Ansprüche

    1. Verfahren für den Betrieb einer Basisstation, BS, (102) in einem Funkkommunikationssystem, umfassend:

    Empfangen (2001) einer Mehrzahl von Signalen, die Informationen zum Beamforming an eine Mehrzahl von Teilnehmergeräten, UE, aufweisen, unter Verwendung eines Full-Dimensional Multiple-Input Multiple-Output, FD-MIMO, Beamforming-Schemas über mindestens eine Zeitressource und mindestens eine Frequenzressource, mit Co-Scheduling an die Mehrzahl von UE;

    Durchführen (506, 1700, 2003) einer Kalibrierung zur Erzeugung eines Vorcodierers auf der Basis der empfangenen Mehrzahl von Signalen, wobei das Durchführen der Kalibrierung folgendes umfasst:

    Erkennen (1704) einer Abtastwertdiskrepanz für eine Mehrzahl von Common Public Radio Interface, CPRI, Verbindungen, die mindestens einem Antennenarray zugeordnet sind;

    Kompensieren (1704) der erkannten Abtastwertdiskrepanz für die Mehrzahl von CPRI-Verbindungen;

    Erkennen (1706) von Verzögerungsdifferenzen über eine Mehrzahl von Kanälen, nachdem die erkannte Abtastwertdiskrepanz kompensiert worden ist;

    Kompensieren (1706) der erkannten Verzögerungsdifferenzen über die Mehrzahl von Kanälen; und

    Anpassen (1708) der Phasen der Mehrzahl von Kanälen, so dass diese phasengleich sind, nachdem die erkannten Phasendifferenzen kompensiert worden sind; und

    Durchführen (2005) des Beamforming an Signalen an die Mehrzahl von UE auf der Basis eines aus einem Ergebnis der Kalibrierung erhaltenen Vorcodierers.


     
    2. Verfahren nach Anspruch 1,
    wobei die Kalibrierung an einem Funk-Frame ausgeführt wird, um eine Zeitsteuerungsanpassung über die Mehrzahl von Common Public Radio Interface-, CPRI, Verbindungen (308) bereitzustellen, wobei jede der Mehrzahl von CPRI-Verbindungen dazu dient, ein HF-Frontend (302) der BS (102) mit einer Basisbandeinheit (306) der BS (102) zu verbinden;
    wobei die Zeitsteuerungsanpassung eine grobe Zeitsteuerungsanpassung umfasst, welche die Abtastwertdiskrepanz erkennt und kompensiert, und eine feine Zeitsteuerungsanpassung, die eine Breitbandkalibrierung erreicht, und wobei das Verfahren ferner folgendes umfasst:

    Durchführen einer Virtualisierung des mindestens einen Antennenarrays gemäß einer zweidimensionalen, 2D, Ebene, die einem von dem Vorcodierer erzeugten Signal zugeordnet ist;

    Aktualisieren des Vorcodierers gemäß einer Schätzung einer Mehrzahl von Kanälen, wobei die Schätzung der Mehrzahl von Kanälen auf der Basis pro Ressourcenblock, RB, durchgeführt wird; und

    Übermitteln der Signale an dem mindestens einen Antennenarray an die Mehrzahl von UE unter Verwendung des FD-MIMO-Beamforming-Schemas.


     
    3. Verfahren nach Anspruch 2, wobei das Verfahren ferner folgendes umfasst:

    Erhalten (1315) eines Einzelbenutzer-, SU, Kanalgüteindikators, CQI, auf der Basis eines von einem UE der Mehrzahl von UE empfangenen Feedbackberichts und eines geschätzten SRS-Kanals;

    Erhalten (1320) eines Mehrfachbenutzer-, MU,CQI auf der Basis des SU CQI, der Vorcodierungs- und der Scheduling-Informationen; und

    Identifizieren (1330) eines angepassten CQI und eines Wertes eines Modulations- und Codierungsschemas, MCS, entsprechend dem angepassten CQI auf der Basis des MU CQI.


     
    4. Verfahren nach Anspruch 2, wobei das Durchführen der Virtualisierung des mindestens einen Antennenarrays folgendes umfasst: Durchführen einer Antennenvirtualisierungs-Vorcodierung unter Verwendung eines Vorcodierungsschemas mit geringer Komplexität, das das K-Fache der Matrixinversion zum 1-fachen reduziert und eine NxN Dimension der Matrixinversion zu KxK, wobei das Vorcodierungsschema mit geringer Komplexität bestimmt wird gemäß einer Kanalmatrix, die näherungsweise bestimmt wird durch einen Vorcodierungsmatrixindikator, PMI, und einen Rauschabstand, SINR, der nach Mapping eines CQI-Feedbacks erhalten wird.
     
    5. Basisstation, BS, (102) in einem Funkkommunikationssystem, wobei die BS folgendes umfasst:

    einen Transceiver; und

    mindestens einen Prozessor, der funktionsfähig mit dem mindestens einen Transceiver gekoppelt ist,

    wobei der mindestens eine Prozessor so gestaltet ist, dass er für folgende Zwecke steuert:

    Empfangen (2001) einer Mehrzahl von Signalen, die Informationen zum Beamforming an eine Mehrzahl von Teilnehmergeräten, UE, aufweisen, unter Verwendung eines Full-Dimensional Multiple-Input Multiple-Output, FD-MIMO, Beamforming-Schemas über mindestens eine Zeitressource und mindestens eine Frequenzressource, mit Co-Scheduling an die Mehrzahl von UE;

    Durchführen (506, 1700, 2003) einer Kalibrierung zur Erzeugung eines Vorcodierers auf der Basis der empfangenen Mehrzahl von Signalen,

    wobei der mindestens eine Prozessor für das Durchführen der Kalibrierung so gestaltet ist, dass er für folgende Zwecke steuert:

    Erkennen (1704) einer Abtastwertdiskrepanz für eine Mehrzahl von Common Public Radio Interface, CPRI, Verbindungen, die mindestens einem Antennenarray zugeordnet sind;

    Kompensieren (1704) der erkannten Abtastwertdiskrepanz für die Mehrzahl von CPRI-Verbindungen;

    Erkennen (1706) von Verzögerungsdifferenzen über eine Mehrzahl von Kanälen, nachdem die erkannte Abtastwertdiskrepanz kompensiert worden ist;

    Kompensieren (1706) der erkannten Verzögerungsdifferenzen über die Mehrzahl von Kanälen; und

    Anpassen (1708) der Phasen der Mehrzahl von Kanälen, so dass diese phasengleich sind, nachdem die erkannten Phasendifferenzen kompensiert worden sind; und

    wobei der Prozessor ferner so gestaltet ist, dass er das Durchführen (2005) des Beamforming an Signalen an die Mehrzahl von UE auf der Basis eines aus einem Ergebnis der Kalibrierung erhaltenen Vorcodierers steuert.


     
    6. BS nach Anspruch 5, wobei: die Kalibrierung an einem Funk-Frame ausgeführt wird, um eine Zeitsteuerungsanpassung über die Mehrzahl von Common Public Radio Interface-, CPRI, Verbindungen (308) bereitzustellen, wobei jede der Mehrzahl von CPRI-Verbindungen dazu dient, ein HF-Frontend (302) der BS (102) mit einer Basisbandeinheit (306) der BS (102) zu verbinden;
    wobei die Zeitsteuerungsanpassung eine grobe Zeitsteuerungsanpassung umfasst, welche die Abtastwertdiskrepanz erkennt und kompensiert, und eine feine Zeitsteuerungsanpassung, die eine Breitbandkalibrierung erreicht, und
    wobei das der mindestens eine Prozessor ferner so gestaltet ist, dass er für folgende Zwecke steuert:

    Durchführen einer Virtualisierung des mindestens einen Antennenarrays gemäß einer zweidimensionalen, 2D, Ebene, die einem von dem Vorcodierer erzeugten Signal zugeordnet ist;

    Aktualisieren des Vorcodierers gemäß einer Schätzung einer Mehrzahl von Kanälen, wobei die Schätzung der Mehrzahl von Kanälen auf der Basis pro Ressourcenblock, RB, durchgeführt wird; und

    Übermitteln der Signale an dem mindestens einen Antennenarray an die Mehrzahl von UE unter Verwendung des FD-MIMO-Beamforming-Schemas.


     
    7. BS nach Anspruch 6, wobei der mindestens eine Prozessor so gestaltet ist, dass er für folgende Zwecke steuert:

    Erhalten (1315) eines Einzelbenutzer-, SU, Kanalgüteindikators, CQI, auf der Basis eines von einem UE der Mehrzahl von UE empfangenen Feedbackberichts und eines geschätzten SRS-Kanals;

    Erhalten (1320) eines Mehrfachbenutzer-, MU,CQI auf der Basis des SU CQI, der Vorcodierungs- und der Scheduling-Informationen; und

    Identifizieren (1330) eines angepassten CQI und eines Wertes eines Modulations- und Codierungsschemas, MCS, entsprechend dem angepassten CQI auf der Basis des MU CQI.


     
    8. Verfahren nach Anspruch 1 oder BS nach Anspruch 5, wobei das mindestens eine Antennenarray der BS mindestens ein Virtualisierungsmuster umfasst, das mindestens ein Zeitdomänensymbol und mehrere virtualisierte Symbole aufweist, und
    wobei das mindestens eine Antennenarray mindestens ein Antennenelement der BS aktiviert.
     
    9. Verfahren nach Anspruch 1 oder BS nach Anspruch 5, wobei eine Scrambling Identifikation, SCID, in mindestens einen Port eines Demodulationsreferenzsignals, DMRS, zugewiesen wird, der auf die Mehrzahl von UE mit Co-Scheduling abgebildet wird, die sich mindestens eine Zeitressource und mindestens eine Frequenzressource teilen.
     
    10. BS nach Anspruch 6, wobei der mindestens eine Prozessor zur Durchführung der Virtualisierung des mindestens einen Antennenarrays so gestaltet ist, dass er für folgenden Zweck steuert.
    Durchführen einer Antennenvirtualisierungs-Vorcodierung unter Verwendung eines Vorcodierungsschemas mit geringer Komplexität, das das K-Fache der Matrixinversion zum 1-fachen reduziert und eine NxN Dimension der Matrixinversion zu KxK, wobei das Vorcodierungsschema mit geringer Komplexität bestimmt wird gemäß einer Kanalmatrix, die näherungsweise bestimmt wird durch einen Vorcodierungsmatrixindikator, PMI, und einen Rauschabstand, SINR, der nach Mapping eines CQI-Feedbacks erhalten wird.
     
    11. Verfahren nach Anspruch 4 oder BS nach Anspruch 10, wobei K bestimmt ist als eine Anzahl bedienter UE, und wobei N bestimmt ist als eine Anzahl von Antennen zur Verwendung für die Übermittlung von Antennenstrahlen an die UE.
     
    12. Verfahren nach Anspruch 1 oder BS nach Anspruch 5, wobei das mindestens eine Antennenarray der BS eine Mehrzahl vertikal angeordneter Felder umfasst, von denen jedes eine Mehrzahl von Subarrays aufweist, die in einer Konfiguration einer Anzahl n horizontal x 1 vertikal angeordnet ist, wobei jedes der Subarrays eine Mehrzahl von Patchelementen aufweist, die mit einem Corporate Feed Network gespeist werden.
     


    Revendications

    1. Procédé de fonctionnement d'une station de base, BS, (102) dans un système de communication sans fil, le procédé comprenant les étapes suivantes :

    réception (2001) d'une pluralité de signaux comprenant des informations pour la formation de faisceaux vers une pluralité d'équipements utilisateurs, UE, en utilisant un schéma de formation de faisceaux de pleines dimensions à entrées multiples et sorties multiples, FD-MIMO, à travers au moins une ressource temporelle et au moins une ressource de fréquence qui sont co-planifiées sur la pluralité d'UE ;

    exécution (506, 1700, 2003) d'un étalonnage pour générer un précodeur basé sur la pluralité de signaux reçus, l'exécution de l'étalonnage comprenant les étapes suivantes :

    détection (1704) d'un décalage de niveau d'échantillonnage pour une pluralité de connexions d'interface radio publique commune, CPRI, associées à au moins un réseau d'antennes ;

    compensation (1704) du décalage de niveau d'échantillonnage détecté pour la pluralité de connexions CPRI ;

    détection (1706) des différences de retard sur une pluralité de canaux après que le décalage de niveau d'échantillonnage détecté a été compensé ;

    compensation (1706) des différences de retard détectées sur la pluralité de canaux ; et

    ajustement (1708) des phases de la pluralité de canaux pour qu'elles soient en phase après que les différences de retard détectées ont été compensées ; et

    exécution (2005) de la formation de faisceau sur des signaux vers la pluralité d'UE sur la base d'un précodeur obtenu à partir d'un résultat de l'étalonnage.


     
    2. Procédé selon la revendication 1,
    l'étalonnage étant effectué sur une trame radio pour fournir un alignement de synchronisation à travers la pluralité de connexions (308) d'interface radio publique commune, CPRI, chacune de la pluralité de connexions CPRI étant destinée à connecter un frontal RF (302) de la BS (102) à une unité de bande de base (306) de la BS (102),
    l'alignement de synchronisation comprenant un alignement de synchronisation grossier détectant et compensant le décalage de niveau d'échantillonnage, et un alignement de synchronisation fin réalisant un étalonnage à large bande, et le procédé comprenant en outre les étapes suivantes :

    exécution d'une virtualisation de l'au moins un réseau d'antennes selon un plan bidimensionnel, 2D, associé à un signal généré par le précodeur ;

    mise à jour du précodeur conformément à une estimation d'une pluralité de canaux, l'estimation de la pluralité de canaux étant effectuée sur une base par bloc de ressources, RB ; et

    transmission des signaux au niveau de l'au moins un réseau d'antennes à la pluralité d'UE en utilisant le schéma de formation de faisceau FD-MIMO.


     
    3. Procédé selon la revendication 2, le procédé comprenant en outre les étapes suivantes :

    obtention (1315) d'un indicateur de qualité de canal, CQI, pour un seul utilisateur, SU, sur la base d'un rapport de rétroaction reçu d'un UE parmi la pluralité d'UE et d'un canal SRS estimé ;

    obtention (1320) d'un CQI pour un utilisateur multiple, MU, sur la base du SU CQI, du précodage et des informations de programmation ; et

    identification (1330) d'un CQI ajusté et d'un niveau de schéma de modulation et de codage, MCS, correspondant au CQI ajusté sur la base du MU CQI.


     
    4. Procédé selon la revendication 2, l'exécution de la virtualisation d'au moins un réseau d'antennes comprenant l'étape suivante : exécution d'un précodage de virtualisation d'antenne en utilisant un schéma de précodage de faible complexité qui réduit K fois l'inversion de matrice à 1 fois et une dimension NxN de l'inversion de matrice à KxK, le schéma de précodage de faible complexité étant déterminé conformément à une matrice de canal qui est approchée par un indicateur de matrice de précodage, PMI, et un rapport signal sur bruit d'interférence, SINR, qui est obtenu après cartographie d'une rétroaction de CQI.
     
    5. Station de base, BS (102) dans un système de communication sans fil, la BS comprenant :

    au moins un émetteur ; et

    au moins un processeur couplé fonctionnellement à l'au moins un émetteur,

    l'au moins un processeur étant conçu pour commander :

    la réception (2001) d'une pluralité de signaux comprenant des informations pour la formation de faisceaux vers une pluralité d'équipements utilisateurs, UE, en utilisant un schéma de formation de faisceaux de pleines dimensions à entrées multiples et sorties multiples, FD-MIMO, à travers au moins une ressource temporelle et au moins une ressource de fréquence qui sont co-planifiées sur la pluralité d'UE, et

    l'exécution (506, 1700, 2003) d'un étalonnage pour générer un précodeur basé sur la pluralité de signaux reçus,

    pour exécuter l'étalonnage, l'au moins un processeur étant conçu pour commander :

    la détection (1704) d'un décalage de niveau d'échantillonnage pour une pluralité de connexions d'interface radio publique commune, CPRI, associées à au moins un réseau d'antennes,

    la compensation (1704) du décalage de niveau d'échantillonnage détecté pour la pluralité de connexions CPRI,

    la détection (1706) des différences de retard sur une pluralité de canaux après que le décalage de niveau d'échantillonnage détecté a été compensé,

    la compensation (1706) des différences de retard détectées sur la pluralité de canaux, et

    l'ajustement (1708) des phases de la pluralité de canaux pour qu'elles soient en phase après que les différences de retard détectées ont été compensées, et

    l'au moins un processeur étant en outre conçu pour commander l'exécution (2005) de la formation de faisceau sur des signaux vers la pluralité d'UE sur la base d'un précodeur obtenu à partir d'un résultat de l'étalonnage.


     
    6. BS selon la revendication 5, l'étalonnage étant effectué sur une trame radio pour fournir un alignement de synchronisation à travers la pluralité de connexions (308) d'interface radio publique commune, CPRI, chacune de la pluralité de connexions CPRI pour connecter un frontal RF (302) de la BS (102) à une unité de bande de base (306) de la BS (102),
    l'alignement de synchronisation comprenant un alignement de synchronisation grossier détectant et compensant le décalage de niveau d'échantillon, et un alignement de synchronisation fin réalisant un étalonnage à large bande, et
    l'au moins un processeur étant en outre conçu pour commander : l'exécution d'une virtualisation de l'au moins un réseau d'antennes selon un plan bidimensionnel, en 2D, associé à un signal généré par le précodeur ;
    la mise à jour du précodeur conformément à une estimation d'une pluralité de canaux, l'estimation de la pluralité de canaux étant effectuée sur une base par bloc de ressources, RB ; et
    la transmission des signaux au niveau de l'au moins un réseau d'antennes à la pluralité d'UE en utilisant le schéma de formation de faisceau FD-MIMO.
     
    7. BS selon la revendication 6, l'au moins un processeur étant conçu pour commander :

    l'obtention (1315) d'un indicateur de qualité de canal, CQI, pour un seul utilisateur, SU, sur la base d'un rapport de rétroaction reçu d'un UE parmi la pluralité d'UE et d'un canal SRS estimé,

    l'obtention (1320) d'un CQI pour un utilisateur multiple, MU, sur la base du SU CQI, du précodage et des informations de programmation ; et

    l'identification (1330) d'un CQI ajusté et d'un niveau de schéma de codage de modulation, MCS, correspondant au CQI ajusté sur la base du MU CQI.


     
    8. Procédé selon la revendication 1 ou BS selon la revendication 5, l'au moins un réseau d'antennes de la station de base comprenant au moins un motif de virtualisation comprenant au moins un symbole dans le domaine temporel et de multiples symboles virtualisés, et
    l'au moins réseau d'antennes activant au moins un élément d'antenne de la BS.
     
    9. Procédé selon la revendication 1 ou BS selon la revendication 5, une identification de brouillage, SCID, étant attribuée à au moins un port de signal de référence de démodulation, DMRS, qui est mis en correspondance avec la pluralité d'UE co-régulés partageant l'au moins une ressource temporelle et l'au moins une ressource de fréquence.
     
    10. BS selon la revendication 6, afin d'exécuter la virtualisation de l'au moins un réseau d'antennes, l'au moins un processeur étant conçu pour commander :
    l'exécution d'un précodage de virtualisation d'antenne en utilisant un schéma de précodage de faible complexité qui réduit K fois l'inversion de matrice à 1 fois et une dimension NxN de l'inversion de matrice à KxK, le schéma de précodage de faible complexité étant déterminé conformément à une matrice de canal qui est approchée par un indicateur de matrice de précodage, PMI, et un rapport signal sur bruit d'interférence, SINR, qui est obtenu après cartographie d'une rétroaction de CQI.
     
    11. Procédé selon la revendication 4 ou BS selon la revendication 10, le K étant déterminé comme un nombre d'UE desservies et le N étant déterminé comme un nombre d'antennes à utiliser pour transmettre des faisceaux d'antenne aux UE.
     
    12. Procédé selon la revendication 1 ou BS selon la revendication 5, l'au moins un réseau d'antennes de la BS comprenant une pluralité de panneaux disposés verticalement, chacun d'entre eux comprenant une pluralité de sous-réseaux qui sont disposés dans une configuration de nombre n horizontal x 1 vertical, chacun des sous-réseaux comprenant une pluralité d'éléments de raccordement alimentés par un réseau d'alimentation d'entreprise.
     




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    Cited references

    REFERENCES CITED IN THE DESCRIPTION



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    Patent documents cited in the description