FIELD OF THE INVENTION
[0001] This invention relates to the field of sound controllers, and in particular to an
adaptive feedforward active noise control method.
BACKGROUND TO THE INVENTION
[0002] Noise control, and particularly active noise reduction, has been an objective for
many years, particularly to reduce the ambient noise in airplanes or in industrial
environments. Such systems have generally utilized feeding a canceling sound (referred
to as antinoise hereinafter) in inverse phase to a sound that is to be reduced or
eliminated. Systems have been designed which are comprised of open loop control systems
or closed loop control systems, analog or digital, the antinoise being applied in
a feedback or in a feedforward system.
[0003] Much of the progress in this field has been directed to the control of noise in a
duct, such as an air conditioning duct or an automotive exhaust pipe.
[0004] However, there has also been a need to control noise in an earcup, such as would
be used by a helicopter pilot.
[0005] One of the early noise control systems directed to control of noise in an earcup
is described in U.S. patent 2,972,018, invented by M.E. Hawley et al. This patent
describes the use of a microphone which picks up sound to be canceled, close to the
exterior of an earcup, then amplifies and phase inverts the sound and feeds the resulting
anti-noise to an earphone that applies the antinoise, which arrives with the acoustic
noise to be canceled, to the interior or the earcup. While this system is analog,
and is therefor fast operating, is an open loop control system, and is not adaptive.
The performance of the device is influenced by changes in coupling of the ambient
noise to the ear (e.g., by changes in the fit of the device, by head movement), by
relative movements of the components, and by the stability of the electronic components.
Moreover it operates using vacuum tubes, and so cannot be practically operated on
batteries. Further, due to its size, it is not portable.
[0006] To overcome these problems, practitioners have advanced the state of the art using
closed loop feedforward digital control systems. A simplified view of such a system
is shown in Figure 1. A sound u
k which is to be controlled passes along a duct 1, and is detected by a microphone
3 from which the signal is passed to a control system 5. An electroacoustic transducer
7 is located downstream of the microphone 3, which injects sound into the pipe, in
accordance with a control signal applied by control system 5, e.g., in intensity,
frequency and phase such as to cancel the sound u
k, resulting in the sound ur
k, which desirably can be null.
[0007] A microphone 9 in the duct downstream from the transducer 7 closes the loop by detecting
any residual sound following the cancellation, and returns an error signal to the
control system 5, which responds by modifying the control signal applied to transducer
7 so as to minimize ur
k detected at microphone 9.
[0008] It will be recognized that the control system 5, operating in the digital mode, has
a limitation in speed based on the inherent operating speed of its processor and due
to the sampling rate of the primary signal from microphone 3, the error signal from
microphone 9 and the control signal provided to the electroacoustic transducer 7.
Consequently in practical systems this approach has been limited to applications,
wherein the microphone can be placed far upstream of transducer 7, in order to be
able to sample the arriving signal as early as possible and thus compensate for the
inherent time delay within a digital system.
[0009] Due to the above-described speed limitation, this system has not been able to be
adapted to the control of random noise entering an earcup without substantial loss
in peformance, since microphone 3 is required to be close to the outside boundary
of the earcup, and therefore there is insufficient processing time available for the
control system to properly control the sound in the earcup.
[0010] Relevant prior art is disclosed in EP-A-471290.
SUMMARY OF THE INVENTION
[0011] Embodiments of the present invention provide means and methods for utilizing a feedforward
digital control system in applications in which the difference in time of arrival
of sound at microphones 3 and 9 is small, such as an earcup sound control apparatus,
with substantial sound control. Key aspects of the invention substantially overcome
the inherent time delay within the control system and reduce the processing load of
the control system, thus allowing it to generate a practical anti-noise control signal
for an earcup type system.
[0012] In accordance with the present invention, we provide a method of noise control of
an acoustic signal comprising:
(a) obtaining a reference signal of the acoustic signal to be controlled,
(b) applying an antinoise signal to the acoustic signal so as to control the acoustic
signal,
(c) obtaining an error signal resulting from the application of the antinoise signal
to the acoustic signal,
(d) generating said antinoise signal from said reference signal by passing the reference
signal through a first filter having controllable filter coefficients,
(e) controlling the filter coefficients by processing the error signal and a modified
representation of the reference signal and generating a coefficient control signal
such as to generate the antinoise signal,
(f) applying the coefficient control signal to the first filter, and
(g) oversampling the reference and error signals, and controlling the first filter
coefficients, and generating the antinoise signal, by processing only a decimated
fraction of the oversampled samples of the reference and error signals, said fraction
being one quarter or less, and applying said antinoise signal to the acoustic signal
at said oversampled rate.
BRIEF INTRODUCTION TO THE DRAWINGS
[0013] A better understanding of the invention will be obtained by considering the detailed
description below, with reference to the following drawings, in which:
Figure 1 is a simplified schematic diagram of a prior art type of feedforward noise
control system,
Figure 2 is a sectional view of an earcup type of noise control system which can be
used in conjunction with the present invention,
Figure 3 is a reproduction of a computer simulation of an adaptive sound controller
from the prior art,
Figure 4 is a block diagram of an embodiment of the present invention,
Figures 5A and 5B illustrate a complete error path impulse response model as in the
prior art, and a simplified error path impulse response model as in an embodiment
of the present invention, respectively,
Figure 6 is a graph illustrating noise reduction without, and with a simplified error
model in accordance with an embodiment of the invention,
Figure 7 is a block diagram of a more detailed structural embodiment of the invention,
Figure 8 is a timing chart used to describe another embodiment of the present invention,
Figure 9 is a schematic diagram of a low order high pass filter that can be used in
an embodiment of the present invention,
Figure 10 illustrates an acoustic filter, in accordance with an embodiment of the
invention,
Figure 11 is a schematic diagram of fixed-ratio gain amplifiers in accordance with
another embodiment of the invention, and
Figure 12 is a graph showing the error path frequency response for the earcup device
having an impulse response shown in Figure 5B, and as measured when the earcup is
poorly sealed to the head.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
[0014] Turning to Figure 2, an earcup type of sound control system is shown. A sound attenuating
earcup 11 is fitted over the ear 13 of a user, and seals to the skin of the user.
A reference microphone 15 is located just outside of the earcup, e.g. on an axis with
the ear canal 17 of the user.
[0015] An electroacoustic transducer (earphone 19) is located within the volume between
the earcup 11 and the ear 13, also preferably on the axis with the ear canal and microphone
15. An error microphone 21 is also located in the volume between the earcup 11 and
the ear 13, also preferably on the aforenoted axis.
[0016] The microphone 15 corresponds to microphone 3 in Figure 1, the earphone 19 corresponds
to the transducer 7 of Figure 1, and the microphone 21 corresponds to the microphone
9 of Figure 1. However, it will be recognized that due to the earcup structure (which
typically may be in the form of ear protectors/earphones of a helicopter pilot), the
microphones 15 and 21 are very close to the earphone 19, allowing little processing
time in the control system due to the very short time that it takes for sound to traverse
these short distances, and it being not practical to move microphone 15 a significant
distance from the earcup.
[0017] The article "Active Adaptive Sound Control In A Duct: A Computer Simulation", by
J.C. Burgess, in the Journal of The Acoustical Society Of America, 70(3), September
1981, pp. 715 - 719, describes a theoretical and computer simulated digital adaptive
controller for the system shown schematically in Figure 1. A schematic of a digital
feedforward sound control system described in that article, is reproduced in Figure
3 herein, from Figure 6 of the article. A description of its operation is considered
redundant herein, as its function will be readily deduced from the aforenoted article
as well as from the description of an embodiment of the invention as shown in Figure
4 below.
[0018] Figure 4 illustrates an embodiment of the present invention. A transmission path
formed by the earcup and cushion against the skin, and air leaks around the cushion,
is represented by H
p. Figure 3, by comparison, shows an air conduction path.
[0019] In Figure 4, the transfer function for the digitized signal derived from the sound
field outside the ear cup, X, is H
1, while that for the error signal, E, is H
3. The corresponding function for the secondary source path is H
2. The corrective sound pressure generated from signal U experiences a further transmission
path in propagating from the earphone to the error microphone, H
a.
[0020] It may be seen that H
2 and H
3 have corresponding transfer function blocks in Figure 3, while the transfer function
H
1 in Figure 4 has corresponding element Δ in Figure 3.
[0021] The signal derived from H
1 (Δ) is applied to an adaptive FIR filter W, which applies an antinoise signal via
transfer function H
2 to a summer Σ, to which the acoustic signal to be controlled is also applied. In
the system of Figure 3, the summer is actually the cavity in front of and in the region
of the transducer 7 within the duct 1, while in the system of Figure 4, the summer
is the region within the cup 11 particularly between the earphone 19 and the ear canal
17. Here the antinoise signal from H
2 output from FIR filter W, passing via transfer functions H
2 (and H
a in Figure 4), is added to the acoustic signal so as to cancel it.
[0022] The coefficients of the filter W are controlled by control system WF (Figure 3),
or control system LMS (Figure 4). These control systems obtain the error signal from
transfer function H
3, represented by E in Figure 4 and e
k in Figure 3, as well as a reference signal R (Figure 4) or v
k (Figure 3). This reference signal is derived by a modification of the sampled reference
signal from microphone 3 (Figure 1), using transfer function H
4 in Figure 3, which forms an error model of the system.
[0023] The error model in the prior art system (Figure 3) is derived from continuous sampling
of the system signals, and is a characteristic of the system. The control system WF,
after an error model has been determined, varies the coefficients of the adaptive
FIR filter W so as to cause an output signal to be applied to the summer to control
the sound which is detected at the error microphone 9.
[0024] A representation of the error path impulse response model of the prior art system
of Figure 3 is illustrated in Figure 5A, wherein each dot on the graph represents
a ditferent sampling time (the horizontal axis representing the time sequence of consecutive
samples).
[0025] Each point is required to be calculated by a processor in control system W, in order
to obtain system identification (characterization). Due to the computational load
on the control system processor, it has been found that it is impractical to operate
a digital feedforward active noise control system wherein there is little time between
the reference sound pickup (microphone 15), the cancellation sound (earphone 19),
and the error sound pickup (microphone 21), as noted earlier.
[0026] In accordance with an embodiment of the present invention the impulse response model
H
e is synthesized so that it eliminates the need for system identification. In a preferred
embodiment of the model H
e, the magnitudes of the FIF filter coefficients, h
i, satisfy the condition:
where i=1,2,3,....N, and N is the total number of filter coefficients.
[0027] A synthesized impulse model in which this condition is satisfied is illustrated in
Figure 5B. It may be seen (by counting the sampling points) that there are 9 computational
points after which all h
i = O. This, in this example N may be set to 9. From the time that the model reaches
0, the computational load on the system processor approaches zero. Compare this with
the prior art system of Figure 5A in which there are 100 non-zero computational points
of which about 34 are visibly non-zero points, which represents a significant load
on the processor, since a value for R (the response) has to be obtained for each recalculation
of the control filter coefficients.
[0028] Figure 6 illustrates the measured active noise reduction of band-limited white noise
using a 200 tap FIR filter W using a true error path model with 200 coefficients h
i (the dashed graph) as in the prior art, as compared with the simplified error model
with 9 coefficients h
i satisfying equation (1), that is N=9, as in the present invention (the solid line).
It may be seen that there is little difference in noise reduction between that obtained
with the true error path model and the simplified error path model.
[0029] Instead of a synthesized error-path impulse response ' model used for H
e, a truncated measured impulse response model, or a truncated synthesized impulse
response model could be used.
[0030] Due to the fact that the coefficients of the error path impulse response average
and rapidly converge to zero, it is clear that the processing load is significantly
decreased. This allows more time for other calculations to be performed during a given
time period.
[0031] Apparatus to implement the present invention is illustrated in Figure 7. The elements
of the cup system shown in Figure 2 are reproduced.
[0032] The outputs X and E of the microphones 15 and 21 respectively are applied to low
pass filters 23A and 23B respectively, in which the bandwidth is limited to low frequencies,
which are the frequencies most likely to penetrate the ear cup. The outputs of the
filters are applied to A/D converters 25A and 25B respectively, in which the analog
signals are converted to digital signals.
[0033] The output signals of A/D converters 25A and 25B are subjected to an interface delay
27A and 27B, and from the interface delays the signals are filtered in decimation
filters 29A and 29B. The interface delays 27A, 27B and 39 are dependent on the hardware
implementations of the active noise control system, which is taken to include any
phase delay in the low pass filters 23A, 23B and 43, and is commonly related to the
sampling time interval.
[0034] The filtered signal from the reference microphone is then applied to error path FIR
filter 31 and to controller FIR 33, while the filtered signal from the error microphone
is applied to LMS control filter adapter 35.
[0035] Error path FIR filter 31 corresponds to and provides the transfer function H
e in Figure 4, and LMS adapter 35 corresponds to the LMS adapter in Figure 4. Controller
FIR 33 in Figure 7 corresponds to FIR filter W in Figure 4, which in a successful
embodiment was a 200 tap FIR filter, controlled by LMS adapter 35.
[0036] The output signal of filter 33 is applied to an interpolation filter 37, after which
the signal is subjected to an interface delay 39. The signal is then converted to
analog form in D/A converter 41, and the resulting analog signal is applied to low
pass filter and earphone driver 43. The canceling or otherwise acoustic modifying
signal from driver 43 is applied to earphone 19.
[0037] It is preferred that the decimation and interpolation filters, error path FIR filter,
controller FIR filter and LMS controller should all be implemented in a digital signal
processor, such as 32 bit floating point type TMS320C31 manufactured by Texas Instruments
Inc., illustrated in Figure 7 as block 45 contained within the dashed line.
[0038] The synthesized simplified error path impulse response model is implemented in error
path FIR 31, to provide a filtered signal to the adapter 35. The LMS controller algorithm
can follow what is described in the aforenoted article by Burgess or algorithms for
feedforward control described in "Active Noise Control:Algorithms and DSP Implementations"
by S.M. Kuo and D.R. Morgan, Wiley, New York, 1996.
[0039] It is preferred that the signals from either or both of the microphones should be
digitally oversampled. Thus instead of sampling at twice the highest noise frequency
to be controlled, it is preferred to sample at a frequency or at frequencies that
are equal to or greater than five times this frequency. This process usually reduces
the interface time delay, especially when the low-pass filters 23A, 23B and 43 are
readjusted to the higher Nyquist frequency.
[0040] Reference is now made to Figure 8, which illustrates timing, using the oversampling
and decimation filters 29A and 29B. The signal in the top graph shows sampling intervals,
t
IO, of the A/D converters 25A and 25B. The frequency of sampling is at the oversampling
rate described above.
[0041] At the time delayed from the first shown sampling instance, the resulting digital
signal is received by the digital signal processor 45, as illustrated in the second
row of Figure 8. It has been found that not all of the sampled data need be processed;
the input data from time spaced samples can be processed, and the second row of Figure
8 illustrates every fourth sample being processed.
[0042] The third row in Figure 8 illustrates that the processing time for each sample passed
to the DSP 45 is less than one sampling interval at the control system sampling rate,
t
CTRL. However it should also be noted that there is substantial time between the completion
of processing of a sample and the initiation of processing of the next. That time
can be used to process another channel (e.g. for a second ear cup), put to other purposes
such as processing additional samples, employing control filters with a larger number
of filter coefficients, or the DSP can remain substantially idle to reduce electrical
power consumption.
[0043] At any given sampling rate, an increase in the total number of control filter coefficients
permits lower frequencies of noise to be controlled.
[0044] As shown in the fourth row, following the completion of processing, the correction
(antinoise) signal for the earphone 19 is passed to the D/A converter. The same digital
correction signal is applied to the earphone at the oversampled rate until the correction
signal changes at which time a changed correction signal (e.g., corresponding to the
fifth, or ninth, oversampled reference input signal) will be applied to the earphone.
[0045] The total time delay between sampling the input signal and the production of the
correction signal may be seen to be only two oversampling delay time intervals t
IO, which is a substantial decrease from the time if the oversampling and decimation
method is not used. This allows the reference microphone 15 to be placed close to
the earcup, i.e. close to the earphone 19, and makes a practical earcup noise canceling
system possible.
[0046] In a successful embodiment, the oversampling frequency was 40 kHz, and the control
frequency, that is, resulting from the processing of a fraction of the oversampled
samples, was 10 kHz (i.e., every fourth sample was processed). The noise bandwidth
was 150-800 Hz.
[0047] The error and/or reference and/or control signals can be filtered by means of electrical,
acoustical and/or electroacoustic filters, as part of transfer functions H
3, H
1 and H
2. Such a filter is illustrated in Figure 4 as filter 47 in the error signal path,
and it is preferred to be a low order analog filter (i.e. a filter with amplitude
changing with frequency of no more than 12 db/octave), for example the high pass filter
shown in Figure 9. The example electrical filter 47 shown is comprised of a pair of
capacitors in series with one conductor and resistors connected across the pair of
conductors between the capacitors and across the input and output. Electrical and
acoustical filters of this type are well known and their operation need not be described
further herein.
[0048] Filter 47 acts to reduce the system response at frequencies at which noise reduction
is not required. Band limiting can result in improved noise reduction performance
at frequencies at which control is required, reduced power and performance requirements
of the secondary acoustic source (earphone 19), and consequent simplification of hardware.
[0049] Filter 47 and filters 23A, 23B and 23C permit spectrum shaping of the reference and/or
error signals to satisfy predetermined performance requirements, such as psychoacoustic
detection criteria or physiological injury criteria.
[0050] With digital oversampling of the signals from one or both microphones, the low pass
filters 23A, 23B and 43 may be replaced by low-order acoustical or electrical filters
to simplify further the device. An example of a low-order, low-pass acoustical filter
applied to the earphone 19 within an earcup is shown in Figure 10, as cavity 49 containing
exit port 50 (e.g. a small tube) in front of the ear canal 17, coupled to loudspeaker
51 or the equivalent contained in a loudspeaker enclosure 52 via a larger diameter
tube 53, being similar in diameter to the active surface of the loudspeaker (the microphones
15 and 21 not being illustrated).
[0051] It is also preferred to extend the dynamic range of the reference and secondary acoustic
sources in such a way that the error path impulse response remains unchanged, or the
error path impulse response and the ratio of the electronic or electroacoustic gains
of the reference and error microphones remain unchanged. This can be realized by specialized
electronic circuits that simultaneously adjust the electronic amplification of signals
X, U and E such that the product of the electronic amplification of signals E and
U remains constant, or the product of the electronic amplification of signals E and
U, and the ratio X/E remain constant.
[0052] To provide the above, as shown in Figure 7, variable fixed-ratio gain amplifiers
49 can be inserted between the low pass filters 23A and/or 23B, and the following
A/D converters 25A and/or 25B respectively, and a variable reciprocal gain amplifier
between D/A converter 41 and low pass filter/driver 43. In Figure 7, the dashed lines
represent a bypass of the straight through conduction path otherwise shown to accommodate
amplifiers 49. A similar structure is inserted in the other conduction paths as noted
above.
[0053] A variable, reciprocal gain and fixed-ratio gain arrangement can be made by means
of linear amplifiers having automatic gain control signal paths, as for example channels
1, 2 and 3 illustrated in Figure 11.
[0054] The circuit can be implemented as shown in Figure 11 by matched field effect transistors
(FETs) 60 and 61 having their source drain circuits respectively connected between
ground and, for FET 60, the noninverting input of operational amplifier 62, and for
FET 61, the inverting input of operational amplifier 63. The inverting input of amplifier
62 is connected to its output and the inverting input of amplifier 63 is connected
to the output through a resistor 65, which has a value R.
[0055] The gate of FET 60 is connected to a gain control input 67 via resistor 69, and to
its source and drain via resistors 71 and 72. Similarly, the gate of FET 61 is connected
to gain control input 67 via resistor 74 and to-its source and drain via resistors
76 and 77.
[0056] The noninverting input of amplifier 62 is connected to input terminal 78, called
channel 1, carrying the U-signal, via resistor 79, which has similar value as resistor
65. Terminal 78 is connected to ground via a resistor 80. The noninverting input of
amplifier 63 is connected to an input terminal 81, called channel 2, carrying the
E signal, and to ground via resistor 83. Output terminals 85 and 87 carry the output
signals of channels 1 and 2 respectively.
[0057] The amplifier circuit for channel 3, carrying the X signal, is similar to that of
channel 2, except for the value of the feedback resistor around the operational amplifier.
An FET 89 which is matched to FETs 60 and 61 has its source-drain circuit connected
between ground and the inverting input of an operational amplifier 91. The gate of
FET 89 is connected via resistor 93 to gain control input 67, and to its source and
drain via resistors 95 and 96. Feedback resistor 98, which has a value R', is connected
between the output of amplifier 91 and its inverting input.
[0058] The input 100 for channel 3, carrying the signal X, is connected to the non-inverting
input of amplifier 91, and to ground through resistor 102.
[0059] The output of the amplifier 91 is connected to output terminal 104.
[0060] Variable gain is provided by matched FETs to obtain the same value of r
ds. Reciprocal gain amplifiers are obtained by choosing circuit values so that the gain
of channel 1 (e.g. carrying the U-signal) is
and channel 2 (e.g. carrying the E signal).
A fixed ratio between the gains of channels 2 and 3 (the latter e.g. carrying the
X signal) is obtained by using circuit values so that
[0061] A laboratory prototype of the above-described invention has also demonstrated that
it adapts to new conditions, such as when the seal between the cushion of the earcup
is broken, as could occur when the user turns his head. This results from the use
of a synthesized error path model, designed according to equation (1), in which an
air leak comparable to that occurring during poor fit of an earcup on the ear has
been included. For example, the frequency response of the synthesized error path model
with impulse response shown in Figure 5B is given by the solid line in Figure 12.
A measured error path frequency response for the same device when the earcup is poorly
sealed to the head is shown by the dashed line in Figure 12.
[0062] A person understanding this invention may now conceive of alternative structures
and embodiments or variations of the above. All those which fall within the scope
of the claims appended hereto are considered to be part of the present invention.
1. A method of noise control of an acoustic signal comprising:
(a) obtaining a reference signal of the acoustic signal to be controlled,
(b) applying an antinoise signal to the acoustic signal so as to control the acoustic
signal,
(c) obtaining an error signal resulting from the application of the antinoise signal
to the acoustic signal,
(d) generating said antinoise signal from said reference signal by passing the reference
signal through a first filter having controllable filter coefficients,
(e) controlling the filter coefficients by processing the error signal and a modified
representation of the reference signal and generating a coefficient control signal
such as to generate the antinoise signal,
(f) applying the coefficient control signal to the first filter, and
(g) oversampling the reference and error signals, and controlling the first filter
coefficients, and generating the antinoise signal, by processing only a decimated
fraction of the oversampled samples of the reference and error signals, said fraction
being one quarter or less, and applying said antinoise signal to the acoustic signal
at said oversampled rate.
2. A method as defined in claim 1 including using a FIR model of a signal path from a
location of the antinoise signal to a location of the error signal to obtain said
modified representation of the reference signal, in which model the filter coefficients
h
i satisfy the condition:
where i=1,2,3,....N, and N is the total number of filter coefficients.
3. A method as defined in claim 2 including using a simplified model of a signal path
from a location of the antinoise signal to a location of the error signal to obtain
said modified representation of the reference signal.
4. A method as defined in claim 2 in which the impulse response model of the error path
is truncated.
5. A method as defined in claim 2 in which the impulse response model of the error path
is synthesized.
6. A method as defined in claim 1 including low order analog frequency shaping of at
least one of the reference signal and the error signal prior to processing, and the
antinoise signal after processing.
7. A method as defined in claim 6 including low order analog frequency shaping using
low-order low-pass acoustical filters.
8. A method as defined in claim 1 including varying the gain of the paths of the reference
signal (X), the antinoise signal (U) and the error signal (E) such that the error
path impulse remains unchanged.
9. A method as defined in claim 8 including varying the gain of the paths of the reference
signal (X), the antinoise signal (U) and the error signal (E) such that the ratio
of the gains of signals X/E remains unchanged.
10. A method as defined in claim 8 including varying the gain of the paths of the reference
signal (X), the antinoise signal (U) and the error signal (E) such that the product
of the gains of signals E and U remains constant.
11. A method as defined in claim 1 including varying the gain of the paths of the reference
signal (X), the antinoise signal (U) and the error signal (E) such that the product
of the gains of signals E and U, and the ratio of the gains signal X/E, remain constant.
12. A method as defined in claim 1 in which the oversampling frequency is about 40 kHz
or lower.
1. Verfahren zur Lärmsteuerung eines akustischen Signals, mit den folgenden Schritten:
(a) Gewinnen eines Referenzsignals von dem zu steuernden akustischen Signal,
(b) Anwenden eines Gegengeräuschsignals auf das akustische Signal, um das akustische
Signal zu steuern,
(c) Gewinnen eines Fehlersignals, das von der Anwendung des Gegengeräuschsignals auf
das akustische Signal herrührt,
(d) Erzeugen des Gegengeräuschsignals aus dem Referenzsignal, indem das Referenzsignal
durch einen ersten Filter mit steuerbaren Filterkoeffizienten läuft,
(e) Steuern der Filterkoeffizienten durch verarbeiten des Fehlersignals und einer
geänderten Darstellung des Referenzsignals sowie Erzeugen eines Koeffizientensteuersignals,
um das Gegengeräuschsignal zu generieren,
(f) Anwenden des Koeffizientensteuersignals auf den ersten Filter und
(g) Abtasten der Referenz und Fehlersignale sowie Steuern der Koeffizienten des ersten
Filters, Erzeugen des Gegengeräuschsignals durch Verarbeitung lediglich eines dezimierten
Anteils der abgetasteten Signale von Referenz und Fehlersignalen, wobei der Anteil
ein Viertel oder weniger ist, und Anlegen des Gegengeräuschsignals an das akustische
Signal mit der Abtastrate.
2. Verfahren nach Anspruch 1, das ein FIR-Modell eines Signalpfads von einer Stelle des
Gegengeräuschsignals zu einer Stelle des Fehlersignals verwendet, um die geänderte
Darstellung des Referenzsignals zu erhalten, wobei in dem Modell die Filterkoeffizienten
h; die folgende Bedingung erfüllen:
wobei i = 1, 2, 3, ...N und N die Gesamtanzahl der Filterkoeffizienten bezeichnet.
3. Verfahren nach Anspruch 2, das ein vereinfachtes Modell für einen Signalpfad von einem
Ort des Gegengeräuschsignals zu einem Ort des Fehlersignals aufweist, um die geänderte
Darstellung für das Referenzsignal zu erhalten.
4. Verfahren nach Anspruch 2, bei dem das Impulsantwortmodell des Fehlerpfads abgeschnitten
ist.
5. Verfahren nach Anspruch 2, bei dem das Impulsantwortmodell des Fehlerpfads synthetisiert
ist.
6. Verfahren nach Anspruch 1, bei dem eine analoge Frequenzumformung mindestens eines
von beiden einschließt: das Referenzsignal sowie das Fehlersignal vor der Verarbeitung
und das Gegengeräuschsignal nach der Verarbeitung.
7. Verfahren nach Anspruch 6, bei dem eine analoge Frequenzumformung niedriger Ordnung
unter Verwendung von akustischen Tiefpaßfiltern niedriger Ordnung erfolgt.
8. Verfahren nach Anspruch 1, das eine Veränderung der Verstärkung der Pfade des Referenzsignals
(X), des Gegengeräuschsignals (U) und des Fehlersignals (E) aufweist, derart, daß
der Fehlerpfadimpuls unverändert bleibt.
9. Verfahren nach Anspruch 8, das eine Änderung der Verstärkung der Pfade des Referenzsignals
(X), des Gegengeräuschsignals (U) und des Fehlersignals (E) aufweist, derart, daß
das Verstärkungsverhältnis der Signale X/E unverändert bleibt.
10. Verfahren nach Anspruch 8, das eine Änderung der Verstärkung der Pfade des Referenzsignals
(X), des Gegengeräuschsignals (U) und des Fehlersignals (E) aufweist, derart, daß
das Produkt der Verstärkung der Signale E und U konstant bleibt.
11. Verfahren nach Anspruch 1, das eine Änderung der Verstärkung der Pfade des Referenzsignals
(X), des Gegengeräuschsignals (U) und des Fehlersignals (E) aufweist, derart, daß
das Produkt der Verstärkungen der Signale E und U sowie das Verstärkungsverhältnis
für Signal X/E konstant bleiben.
12. Verfahren nach Anspruch 1, dadurch gekennzeichnet, daß die Abtastfrequenz ungefähr 40 kHz oder weniger beträgt.
1. Procédé de régulation de bruit pour un signal acoustique, comprenant les opérations
suivantes :
(a) obtenir un signal de référence du signal acoustique à réguler,
(b) appliquer un signal d'anti-bruit (ou bruit annulant) au signal acoustique afin
de réguler le signal acoustique,
(c) obtenir un signal d'erreur en résultat de l'application du signal d'anti-bruit
au signal acoustique,
(d) produire ledit signal d'antlbruit à partir dudit signal de référence en faisant
passer le signal de référence dans un premier filtre dont les coefficients sont ajustables,
(e) ajuster les coefficients du filtre en traitant le signal d'erreur et une représentation
modifiée du signal de référence et produire un signal d'ajustement de coefficients
de façon à produire le signal d'anti-bruit,
(f) appliquer le signal d'ajustement des coefficients au premier filtre, et
(g) suréchantillonner le signal de référence et le signal d'erreur et ajuster les
coefficients du premier filtre, puis produire le signal d'anti-bruit, en ne traitant
qu'une fraction décimée des échantillons suréchantillonnés des signaux de référence
et d'erreur, ladite fraction étant d'un quart ou moins, et en appliquant ledit signal
d'anti-bruit au signal acoustique à la fréquence dudit suréchantillonnage.
2. Procédé selon la revendication 1, comportant l'utilisation d'un modèle à réponse impulsionnelle
finie, noté modèle FIR, d'un trajet de signal qui part de l'emplacement du signal
d'anti-bruit pour aller à l'emplacement du signal d'erreur, afin d'obtenir ladite
représentation modifiée du signal de référence, dans lequel modèle les coefficients
de filter h
l satisfont la condition :
où i=1, 2, 3, ... N, et où N est le nombre total de coefficients de filtre.
3. Procédé selon la revendication 2, comportant l'utilisation d'un modèle simplifié d'un
trajet de signal qui part de l'emplacement du signal d'anti-bruit pour aller à l'emplacement
du signal d'erreur afin d'obtenir ladite représentation modifiée du signal de référence.
4. Procédé selon la revendication 2, où le modèle à réponse impulsionnelle du trajet
d'erreur est tronqué.
5. Procédé selon la revendication 2, où le modèle à réponse impulsionnelle du trajet
d'erreur est synthétisé.
6. Procédé selon la revendication 1, comportant une conformation de fréquence analogique
d'ordre Inférieur d'au moins un signal pris entre le signal de référence et le signal
d'erreur avant le traitement, et du signal d'anti-bruit après le traitement.
7. Procédé selon la revendication 6, comportant une conformation de fréquence analogique
d'ordre inférieur qui utilise des filtres acoustiques passe-bas d'ordre inférieur.
8. Procédé selon la revendication 1, comportant l'opération qui consiste à faire varier
le gain des trajets du signal de référence (X), du signal d'anti-bruit (U) et du signal
d'erreur (E) de façon que l'impulsion du trajet d'erreur reste inchangée.
9. Procédé selon la revendication 8, comportant l'opération qui consiste à faire varier
le gain des trajets du signal de référence (X), du signal d'anti-bruit (U) et du signal
d'erreur (E) de façon que le rapport des gains des signaux X/E reste inchangé.
10. Procédé selon la revendication 8, comportant l'opération qui consiste à faire varier
le gain des trajets du signal de référence (X), du signal d'anti-bruit (U) et du signal
d'erreur (E) de façon que le produit des gains des signaux E et U reste inchangé.
11. Procédé selon la revendication 1, comportant l'opération qui consiste à faire varier
le gain des trajets du signal de référence (X), du signal d'anti-bruit (U) et du signal
d'erreur (E) de façon que le produit des gains des signaux E et U et le rapport des
gains des signaux X/E restent inchangés.
12. Procédé selon la revendication 1, où la fréquence de suréchantillonnage est d'environ
40 kHz ou moins.