Field of the Invention
[0001] The present invention relates to a Radio Frequency reflection type phase shifter,
and a method of Radio Frequency reflection type phase shifting.
[0002] Some background is provided by United Kingdom patent
GB186541A and United States patent
US6172385B1.
Summary
[0003] The reader is referred to the appended independent claims. Some preferred features
are laid out in the dependent claims.
[0004] An example of the present invention is a Radio Frequency reflection type phase shifter,
the phase shifter comprising a coupler for input and output, and N variable capacitors,
where N is an integer value of 2 or more, each of the variable capacitors providing
radio frequency reflection, each of the variable capacitors being connected to the
coupler by at least one of the impedance transformers, the characteristic impedances
of the impedance transformers having been selected so that the phase shifter provides
a phase shift at least substantially proportional to the value of N, wherein each
of the variable capacitors comprises electrochromic material.
[0005] The inventors realised that on the one hand capacitors using electrochromic materials
were possible, for example as described in United States Patent Publication
US2015/00325897A1.
[0006] The inventors realised on the other hand a circuit as described in European Patent
Publication
EP2996190 A1 was available using capacitors in the form of varactor diodes.
[0007] The inventors realised that the circuit described in
EP2996190A1 could be adapted to instead use capacitors using electrochromic materials as described
in
US2015/00325897A1 in order to provide a useful and improved phase shifter
[0008] As compared to the prior approach described in
EP2996190A1, four advantages of using EC based material as opposed to varactor diodes as active
elements in the configuration of the proposed phase shifters are: (a) The exact values
of the "ON" and "OFF" capacitance of EC based materials can be tailored by the surface
area of the electrode end pads (this is not possible with varactor diodes); (b) The
capacitance ratio between the "ON" and "OFF" state can be tailored by the appropriate
choice of the electrolyte, for which we have in-house experience; (c) Varactor diodes
exhibit a significant non-linear behaviour, whereas EC based materials are highly
linear; (d) A possibility exists to actuate EC based materials by light, whereas this
is not possible with varactor diodes.
[0009] Preferably each of the variable capacitors comprises an electrolyte element and at
least one electrochromic element between a first electrode and a second electrode.
[0010] Preferably, the first electrode comprises a ground plate on which lies the first
electrochromic element, and the electrolyte element lies on the electrochromic element,
the electrochromic element comprising an electrochromic layer, and the electrolyte
element comprising an electrolyte layer.
[0011] Preferably each of the variable capacitors further comprises a second electrochromic
element between the electrolyte element and second electrode, the second electrochromic
element comprising a second electrochromic layer.
[0012] Preferably the coupler is a 3dB-coupler having four ports, N'/2 of the variable capacitors
being connected to the coupler via one of two of the ports, and N'/2 of the capacitors
being connected to the coupler via a second of said two ports, where N' is an even
number integer of 4 or more. Alternatively preferably the coupler is a circulator
having three ports, the N variable capacitors being connected to the circulator via
one of the ports.
[0013] Preferably the impedance transformers are microstrip lines.
[0014] Preferably the characteristic impedances of the impedance transformers are selected
in according with a selected value of a parameter value q determined for a given capacitor
as
where Z
0 is the characteristic impedance of the impedance transformers, Xmin is the minimum
reactance of the capacitor and Xmax is the maximum reactance of the capacitor.
[0015] Preferably the capacitance of each of the variable capacitors is variable by adjusting
a d.c. voltage applied across the capacitors. Preferably said phase shift is at least
substantially proportional to the value of N when a mid-range value of the capacitance
of the variable capacitors is selected so the corresponding reactance at an operating
radio frequency is the characteristic impedance Z
0. Preferably the capacitors are variable between a higher capacitance 'fully ON' state
when the d.c. voltage is at a first level and a lower capacitance 'OFF' state when
the d.c. voltage is at a second level.
[0016] Some preferred embodiments provide, as compared to existing solutions using EC materials,
greater amounts of phase shift for lower insertion losses. Some preferred embodiments
are suitable for the microwave frequency range.
[0017] Examples of the present invention also relate to corresponding methods.
[0018] An example of the present invention relates to a method of Radio Frequency reflection
type phase shifting, by: applying an input signal to a phase shifter comprising a
coupler for input and output, and N variable capacitors, where N is an integer value
of 2 or more, each of the variable capacitors providing radio frequency reflection,
each of the variable capacitors being connected to the coupler by at least one impedance
transformer, the characteristic impedances of the impedance transformers having been
selected so that the phase shifter provides a phase shift at least substantially proportional
to the value of N, wherein each of the variable capacitors comprises electrochromic
material; and receiving an output signal from the coupler.
[0019] Preferably the capacitance of each of the variable capacitors is variable by adjusting
a d.c. voltage applied across the capacitors.
[0020] Preferably said phase shift is at least substantially proportional to the value of
N when a mid-range value of the capacitance of the variable capacitors is selected
so the corresponding reactance at an operating radio frequency is the characteristic
impedance Z
0.
Brief Description of the Drawings
[0021] Embodiments of the present invention will now be described by way of example and
with reference to the drawings, in which:
Figure 1 is a diagram illustrating a known Radio Frequency (RF) reflective type phase
shifter (PRIOR ART),
Figure 2 is a diagram illustrating a phase shifter which is two of the phase shifters
shown in Figure 1 cascaded (PRIOR ART),
Figure 3 is a diagram illustrating phase shifter which is three of the phase shifters
shown in Figure 1 cascaded (PRIOR ART),
Figure 4 is a general circuit diagram illustrating a generalised circuit of an RF
reflective type phase shifter according to a first embodiment of the invention, generalised
in the sense it is n-th order where n is two or more,
Figure 5 is a circuit diagram illustrating part of the reflective loads portion of
the circuit shown in Figure 4,
Figure 6 is a cross sectional view of a parallel plate capacitor including electrochromic
(EC) material, multiple of which are used in the phase shifter shown in Figures 4,
Figure 7 is a perspective view of the generalised n-th order reflection type phase
shifter shown in Figure 4,
Figure 8 is a diagram illustrates the phase shifter shown in Figures 4 and 7 with
n selected to be three, and
Figure 9 is a diagram illustrating the phase shifter shown in Figures 4 and 7 with
n selected to be two.
Detailed Description
[0022] We will first briefly outline the inventor's understanding of some earlier known
approaches then focus in detail on embodiments of the present invention.
Earlier Known Approaches
[0023] A high frequency phase shifter based on EC materials is known from US Patent Publication
US 2015/0325897A1. This high frequency phase shifter is based on the use of Electochromic (EC) material
as bulk, dc induced tunable media in a circuit.
[0024] The inventors realised that this known high frequency EC material based phase shifter
did not exploit the potential of EC materials. In particular, the circuit in that
phase shifter only allowed modest values of phase shifts, typically up to 15 -30 degrees
at frequencies around 3 GHz. In any particular case, the exact value of the phase
shift obtained is, of course, dependent on the frequency of operation and the type
and thickness of the EC material used, however, there is always a limitation as to
how much phase shift can be obtained. Accordingly, the inventors saw a need for new
architectures for high frequency phase shifters based on EC materials.
[0025] This known approach of
US 2015/0325897A1 is illustrated in Figure 1. As shown in Figure 1, there is a ground plate on which
an electrochromic layer is provided, and input and outputs connected via a 3dB coupler
to microstrip contacts contacting the top of the EC layer.
[0026] The inventors realised that a problem with the phase shifter configuration of Figure
1 lies in its inherently low values of achievable phase shifts.
[0027] The inventors realised that the amount of phase shift from the proposed configuration
could be increased by cascading several structures of Figure 1 as shown in Figures
2 and 3, but such an arrangements has drawbacks of increased structural size and complexity,
and increased losses. As regards increased losses, since the number of 3-dB couplers
will be increased, so will its corresponding insertion loss. The increase of the number
of 3-dB couplers is particularly detrimental, since the radio frequency signal in
any 3-dB coupler travels twice through - first to reach the reflective loads and second,
back to reach the input/output ports. A 3-dB coupler is a radio frequency (RF) device
which splits an input RF signal into two signals equal in magnitude, but with a 90°
phase shift between them.
[0028] From noting these drawbacks, the need for different architectures for high frequency
phase shifters based on EC materials became evident to the inventors.
Example Embodiments
[0029] We now turn to describing some preferred embodiments.
[0030] Firstly, a generic circuit for a range of phase shifters will be described. Secondly,
the active elements used in the generic circuit will described, namely capacitors
using EC material. Thirdly, example reflective type phase shifters will described
that are in accordance with the generic circuit.
[0031] As will be seen, in the generic circuit of Figures 4 and 5, parallel plate capacitors
are used formed using Electrochromic material as shown in Figure 6.
[0032] After this description, we will present some comparison data comparing some properties
of embodiments to examples of known approaches.
Circuit
[0033] The input admittance of the circuit in one of the reflective loads of the proposed
circuit of Fig. 4 is represented as
[0034] Where,
[0035] Or, in general,
[0036] Here, k
i,j, i = 2...n, j = 1, 2 represent the impedance transformers, n represents the order
of the absorptive filter and Y = Z
-1. It can be inferred from (2) - (3) that the input admittance, Y
in, can be represented in the form of a generalized continued fraction
[0037] Or equivalently,
where
a
1 =1,
∀ n ≥ 2,k=1...n-1 and
[0038] The input admittance of the n-th order absorptive transmission zero can now be represented
as
where A
m-1 =b
m-1A
m-2+a
m-1A
m-3 and B
m-1 = b
m-1B
m-2+a
m-1B
m-3.
[0039] Solving (6), one obtains the n-th order admittance polynomial from which the expression
for the n-th order polynomial expression for the transmission coefficient of the notch
filter (with the 3-dB coupler included) can be derived
where
Y0 is the characteristic admittance of the 3-dB coupler.
[0040] Substituting (6) into (7) and converting the admittance parameters into their impedance
counterparts, i.e.
and Y = Z
-1 one obtains the expression for the transmission coefficient,
S21, as a function of impedance parameters
[0041] In order for (8) to offer phase shift increase commensurate with the number of active
elements in the reflective loads, (8) needs to be represented in the following form
where n indicates the number of pairs of active elements in the circuit of the reflective
load, Figs. 4 and 5. More generally, (9) can be written as
where q indicates the position of the transmission zero on the resistance scale which
can be adjusted with a proper selection of the impedance transformers k
i,j,i = 2...n, j = 1, 2. The phase shift provided by (10) can be written as
[0042] For
q=1, the phase shift of the proposed structure of Figs. 4 and 5 is increased n-times.
Nevertheless, simply setting
q=1, does not necessarily result in the optimal phase shift. The optimal phase shift
is found by finding the roots of
yielding the following 6
th order polynomial
[0044] The first four roots of (13) are always complex conjugate, while the remaining two
roots are real with equal magnitude, but opposite signs. As such, there is always
one solution to (13) that yields the optimum value of the parameter
q. The expression given by (14) can be simplified if it can be assumed that the parasitic
resistance of an active element (i.e. parallel plate capacitor including EC material)
can be neglected. This is a valid assumption in most cases, since this resistance
is typically of the order of 1- 2 ohms. By setting
R = 0 (14) the optimal value q becomes
[0045] The insertion loss of the proposed phase shifter (log scale) is
[0046] Here, the first term on the right represents the insertion loss of the reflective
circuit of the proposed phase shifter. For
q=1 the insertion loss of the proposed reflective load is n-times higher than the insertion
loss of the first order reflective circuit, while if parameter
q is set in accordance with (13), the insertion loss of the reflective loads is always
lower than that achieved with
q=1. The second term on the right is the insertion loss of a 3-dB coupler.
[0047] For comparison, a known phase shifter having cascade connection of n first order
circuits will yield the same phase shift as (11), however, its insertion loss will
be
[0048] In quantitative terms, the reduction in the overall insertion loss of the proposed
circuit over the known phase shifter having cascade connection is
[0049] In view of (17) and (18), (11) and (16) demonstrate the potential of the proposed
circuit - to increase the amount of phase shift of the phase shifter in a linear fashion
with respect to the pairs of active elements, without increasing the insertion loss
in the same linear fashion. For example, if the insertion loss of a 3-dB coupler is
0.3 dB (2
∗0.3 dB in the phase shifter configuration), and for n=2,
q=1 the reduction of the insertion loss using the proposed circuit over the conventional
cascade connection is
[0050] In the derivation of the above equations the condition stipulated in the previous
section related to the retention of a minimum number of 3-dB couplers in the design
of the phase shifter has been fulfilled.
[0051] The active elements are capacitors formed using EC material as will be described
next below.
Capacitors using EC material
[0052] Figure 6 shows a parallel plate capacitor 10 in cross-section.
[0053] As shown in Figure 6, there is a ground plate 12 on which lies a first electrochromic
layer 14 and a second electrochromic layer 18 separated by an electrolyte layer (in
other words a dielectric layer) 16. On top of the second electrochromic layer is a
top electrode 20. It may be considered that the ground plate (also known as the ground
electrode) 12 and the top electrode 20 effectively "sandwich" the intermediate active
layers 14, 16,18.
[0054] An electrochromic material is a material the optical absorption/transmission characteristics
of which can be reversibly changed by the application of an external voltage, light
source, or electric field. Examples include (i) transition-metal and inorganic oxides
such as tungsten oxide, (ii) small organic molecules such as viologens, and (iii)
polymers such a poly-viologens and derivatives of polythiophence, polypyrrole and
polyaniline.
[0055] The first EC layer 14 comprises a suitable EC material, such as WO
3 in this example. In other examples, the EC material is TiO
2, MoO
3, Ta
2O
5, Nb
2O
5, or another of the above -mentioned electrochromic materials.
[0056] The second EC layer 18 comprises NiO in this example. In other examples this layer
is Cr
2O
3, MnO
2, FeO
2, CoO
2, RhO
2, IrO
2, or another suitable material. In this example, the second EC layer 18 acts as an
ion-storage layer.
[0057] In operation, the application of a d.c. bias voltage between the ground plate 12
and top electrode 20 induces changes in the dielectric characteristics of the intermediate
layers 14,16,18 and hence their capacitance as a function of the applied d.c. voltage.
In this example, ground plate 12 is a cathode and the top electrode is an anode.
[0058] The electrolyte layer 16 acts as an ion-conductor layer. The electrolyte layer 16
serves as a reservoir of ions for injection into the first EC layer 14. In this example,
the electrolyte layer 16 also receives ions from the second EC layer 18.
[0059] When voltage is applied via electrical leads 22,24, a corresponding electric field
is generated between the ground electrode 12 and top electrode 20. This electric field
causes ions to be introduced into the first EC layer 14 from electrolyte layer 16.
The electric charge caused by this injection of ions into the first EC layer 14 is
neutralised by a corresponding charge balancing counter-flow of electrons from ground
electrode 12.
[0060] In use the voltage is adjustable to vary the capacitance of the capacitor 10 and
is set to provide a capacitance corresponding to an impedance of characteristic impedance
Z
0, where Z
0 is the characteristic impedance of the Figure 4 circuit. More specifically, the mid-range
capacitance C
mid of the capacitor 10 is selected (where C
mid = 0.5
∗(C
max + C
min) so that the reactance of the capacitor, which is a function of radio frequency and
capacitance, matches the characteristic impedance Z
0 of the Figure 4 circuit. In other word, the mid-range capacitance C
mid of the capacitor 10 is selected so that Z
mid = Z
0 where Z
mid=1/(ωC
mid).
[0061] With the EC capacitors set to have this capacitance, the phase shift provided by
the phase shifter is at least substantially proportional to N where N is the number
of reflective loads, in other words the number of capacitors .
Specific Reflective Type Phase Shifter Examples
[0062] A notional phase shifter 30 of nth order is shown in Figure 7 which is in accordance
with the circuit shown in Figure 4 .
[0063] As now shown in Figure 7, and as previously mentioned in relation to Figure 4, k
i,j,i = 2...n, j = 1, 2 represent the impedance transformers, and n represents the order
of the phase shifter which may be considered an absorptive filter.
[0064] As shown in Figure 7, the impedance transformers k
i,j,i = 2...n, j = 1,2 are formed by microstrip lines 26 over a supporting substrate
28. The capacitors 10 are embedded in the substrate such that, each capacitor 10 has
its respective top electrode 20 flush with (in other words in the same plane as) the
top surface of the supporting substrate 28 so that the microstrip lines 26 can run
flat. The microstrip line 26 has portions of different selected widths, hence different
cross-sectional areas, to provide the respective impedance transformers.
[0065] As previously mentioned, as now shown in Figure 7, a 3-dB coupler 32 is a radio frequency
(RF) device which splits an input RF signal into two signals equal in magnitude, but
with a 90° phase shift between them for transmission to the capacitors 10. The 3dB-coupler
has two input/output ports 34 and two other ports 36 for connection to the capacitors
10.
[0066] In some otherwise similar embodiments (not shown), the 3-dB coupler is replaced by
a circulator (not shown). A circulator has three ports (one port less than the 3-dB
coupler). Two ports of the circulator are input/output ports, whereas the last, third
port is the port to which two or more reflective loads are connected. Each reflective
load comprises a variable capacitor comprising EC materials as described with respect
to Figure 6, connected by at least two impedance transformers as described above made
up of portions of microstrip line of different widths.
n=3 Example
[0067] Figure 8 shows a phase shifter where its circuit is as shown in Figures 4 and 7 with
n selected as three. In other words, Figure 8 shows the 3
rd order reflective type phase shifter.
[0068] In one example, let us assume that the capacitance ratio between the "ON" and "OFF"
state of the EC material based capacitors 10 is 2 (C
max/C
min = 2) and that C
min = 0.4 pF and that the EC material formed capacitors 10 have an equivalent parasitic
resistance of 1 ohm.
Calculation of impedance values for the impedance transformers in this n=3 example
[0069] Setting n = 3 in (5) and substituting (5) into (8), the following expression for
the transmission coefficient is obtained
where,
and
The transmission zero condition is achieved by setting
S21 = 0. In this case, a third order polynomial in Z is obtained and needs to be solved
so that it has a multiple and real root. This is accomplished by setting the discriminant,
Δ, to be zero.
[0070] The condition that the discriminant of (21) is zero yields a triple zero at
[0071] Solving (22) one obtains a quart-quadratic equation in
given by
where,
and
The double zero in
is achieved at
with a condition that the discriminant of (23), Δ
1 =B
2 -4AC, disappears. This condition yields a third order polynomial in
given by
where, D=-512 ,
and
The triple zero of (25) is achieved at
provided that the discriminant of (25) disappears. It can be shown that the discriminant
of (32) is always equal to zero, regardless of the value assigned to
This infers that the triple and identical zero of the polynomial given by (25) is
always achieved and that
can be used as a parameter. Substituting (26) into (24), one finds the expression
for
where k
22 and
are used as parameters
[0072] The relationship between k
22 and the rest of impedance transformers is found from (29). Imposing that the triple
zero of (20) occurs at q · Z
0, where q is a parameter that dictates the position of the transmission zero on the
resistance scale, one obtains the following relationship for k
22
[0073] Substituting (28) into (27), the expression for
now becomes
[0074] The following conditions for the characteristic impedances, k
12, k
11 and k
22 can now be expressed as
where Z
0, q and k
21 are used as parameters. Since, Z
0 is usually, but not necessarily, 50Ω, only k
21 and q can be used in the adjustment of the rest of the impedances of the quarter-wave
transformers, k
12, k
11 and k
22.
n=2 Example
[0075] Figure 9 shows a phase shifter where its circuit is as shown in Figures 4 and 7 with
n selected as two. In other words, Figure 9 shows the 2nd order reflective type phase
shifter.
[0076] In one example, let us assume that the capacitance ratio between the "ON" and "OFF"
state of the EC material based capacitors 10 is 2 (C
max/C
min = 2) and that C
min = 0.4 pF and that the EC material formed capacitors 10 have an equivalent parasitic
resistance of 1 ohm.
Calculation of impedance values for the impedance transformers in this n=2 example
[0077] Setting n = 2 in (5) and substituting (5) into (8), the following expression for
the transmission coefficient is obtained
[0078] (30) assumes that the 3-dB coupler is ideal. The zeroes of (30) yield the following
values for the transformers k
11 and k
12
[0079] Upon which (32) becomes
[0080] By setting q=lit follows
and k
12 =Z
0.
Comparison
[0081] For this comparison, with prior art approaches involving capacitors using EC materials,
it is assumed that the capacitance ratio between the "ON" and "OFF" state of the EC
material based capacitors 10 is 2 (C
max/C
min = 2) and that C
min = 0.4 pF and that the EC material formed capacitors 10 have an equivalent parasitic
resistance of 1 ohm.
[0082] Based on the information on the variable EC material based variable capacitors, the
two phase shifters (one second order and one third order) shown in Figures 8 and 9
were compared against prior art phase shifters involving capacitors using EC materials.
In the design of all these phase shifters, a 3-dB coupler with an insertion loss of
0.3 dB is used. This is a realistic assumption and is evidenced in many practical
designs. All three phase shifters are designed to operate at a centre frequency of
2. 5 GHz. Their performance is summarized in table 1 below.
Table 1 Performance comparison of first, second and third order reflective type phase
shifters
Type of Phase Shifter |
Insertion phase (deg.) |
Insertion loss (dB) |
First order reflective type EC material based phase shifter (Fig. 1 PRIOR ART) |
29.4 |
0.7 |
Second order reflective type EC material based phase shifter (Fig. 9) |
58.6 |
0.8 |
Third order reflective type EC material based phase shifter (Fig. 8) |
88.9 |
0.9 |
Second order reflective type EC material based phase shifter obtained by cascade connection
of two first order phase shifters (Fig. 2 PRIOR ART) |
58.8 |
1.4 |
Third order reflective type EC material based phase shifter obtained by cascade connection
of three first order phase shifters(Fig. 3 PRIOR ART ) |
88.2 |
2.1 |
[0083] As shown in this table the proposed reflective type EC material based phase shifters
of order two or more (n=2,3,4..) offer the benefits of lower loss and increased phase
shift compared to an earlier approach. This can be seen, for example, in comparing
the "second order" data, namely second and fourth rows of data in Table 1. This can
also be seen, for example by comparing the "third order" data, namely the third and
fifth row of data in Table 1.
Some advantages and further details
[0084] As compared to the prior approach described in
EP2996190A1, some advantages of using EC based material as opposed to varactor diodes as the
active elements in the configuration of the proposed phase shifters are as follows.
[0085] First, the exact values of the "ON" and "OFF" capacitance of EC based materials can
be tailored by the surface area of the top electrodes- this is not possible with varactor
diodes.
[0086] Secondly, the capacitance ratio between the "ON" and "OFF" state can be tailored
by the appropriate choice of the electrolyte, for which we have in-house experience.
[0087] Thirdly, varactor diodes exhibit significant non-linear behaviour, whereas EC based
materials are highly linear.
[0088] Fourthly, a possibility exists in some other embodiments to actuate EC based materials
by light, whereas this is not possible with varactor diodes.
[0089] The present invention may be embodied in other specific forms without departing from
its essential characteristics. The described embodiments are to be considered in all
respects only as illustrative and not restrictive. The scope of the invention is,
therefore, indicated by the appended claims rather than by the foregoing description.
All changes that come within the meaning of the claims are to be embraced within their
scope.
[0090] A person skilled in the art would readily recognize that steps of various above-described
methods can be performed by programmed computers. Some embodiments relate to program
storage devices, e.g., digital data storage media, which are machine or computer readable
and encode machine-executable or computer-executable programs of instructions, wherein
said instructions perform some or all of the steps of said above-described methods.
The program storage devices may be, e.g., digital memories, magnetic storage media
such as a magnetic disks and magnetic tapes, hard drives, or optically readable digital
data storage media. Some embodiments involve computers programmed to perform said
steps of the above-described methods.
1. A Radio Frequency reflection type phase shifter (30),
the phase shifter comprising a coupler (32) for input and output (34), and N variable
capacitors (10), where N is an integer value of 2 or more, each of the variable capacitors
providing in use radio frequency reflection,
each of the variable capacitors being connected to the coupler by at least one impedance
transformer (K, 26), the characteristic impedances of the impedance transformers having
been selected so that the phase shifter provides a phase shift at least substantially
proportional to the value of N, characterised in that each of the variable capacitors comprises electrochromic material.
2. A Radio Frequency reflection type phase shifter according to claim 1, in which each
of the variable capacitors (10) comprises an electrolyte element (16) and at least
one electrochromic element (14, 18) between a first electrode (12) and a second electrode
(20)
3. A Radio Frequency reflection type phase shifter according to claim 2, in which the
first electrode (12) comprises a ground plate on which lies the first electrochromic
element (14), and the electrolyte element (16) lies on the electrochromic element
(14), the electrochromic element comprising an electrochromic layer, and the electrolyte
element comprising an electrolyte layer.
4. A Radio Frequency reflection type phase shifter according to claim 2 or claim 3, in
which each of the variable capacitors further comprises a second electrochromic element
(18) between the electrolyte element and second electrode, the second electrochromic
element comprising a second electrochromic layer.
5. A Radio Frequency reflection type phase shifter according to any preceding claim,
in which the coupler (32) is a 3dB-coupler having four ports, N'/2 of the variable
capacitors being connected to the coupler via one of two of the ports, and N'/2 of
the capacitors being connected to the coupler via a second of said two ports, where
N' is an even number integer of 4 or more.
6. A Radio Frequency reflection type phase shifter according to any of claims 1 to 4,
in which the coupler is a circulator having three ports, the N variable capacitors
being connected to the circulator via one of the ports.
7. A Radio Frequency reflection type phase shifter according to any preceding claim,
in which the impedance transformers (K, 26) are microstrip lines.
8. A Radio Frequency reflection type phase shifter according to any preceding claim,
in which the characteristic impedances of the impedance transformers are selected
in accordance with a selected value of a parameter value q determined for a given
capacitor as
where Z
0 is the characteristic impedance of the impedance transformers, Xmin is the minimum
reactance of the capacitor and Xmax is the maximum reactance of the capacitor.
9. A Radio Frequency reflection type phase shifter according to any preceding claim,
in which the capacitance of each of the variable capacitors (10) is variable by adjusting
a d.c. voltage applied across the capacitors.
10. A Radio Frequency reflection type phase shifter according to claim 9, in which said
phase shift is at least substantially proportional to the value of N when a mid-range
value of the capacitance of the variable capacitors (10) is selected so the corresponding
reactance at an operating radio frequency is the characteristic impedance Z0.
11. A Radio Frequency reflection type phase shifter according to claim 10, in which the
capacitors are variable between a higher capacitance ' fully ON' state when the d.c.
voltage is at a first level and a lower capacitance 'OFF' state when the d.c. voltage
is at a second level.
12. A method of Radio Frequency reflection type phase shifting, by:
applying an input signal to a phase shifter (30) comprising a coupler (32) for input
and output, and N variable capacitors (10), where N is an integer value of 2 or more,
each of the variable capacitors providing radio frequency reflection, each of the
variable capacitors being connected to the coupler by at least one impedance transformer,
the characteristic impedances of the impedance transformers (K, 26) having been selected
so that the phase shifter provides a phase shift at least substantially proportional
to the value of N, wherein each of the variable capacitors comprises electrochromic
material; and
receiving an output signal from the coupler.
13. A method of Radio Frequency reflection type phase shifting according to claim 12,
in which the capacitance of each of the variable capacitors (10) is variable by adjusting
a d.c. voltage applied across the capacitors.
14. A method of Radio Frequency reflection type phase shifting according to claim 13,
in which said phase shift is at least substantially proportional to the value of N
when a mid-range value of the capacitance of the variable capacitors (10) is selected
so the corresponding reactance at an operating radio frequency is the characteristic
impedance Z0.
1. Funkfrequenzreflexionstyp-Phasenschieber (30), wobei der Phasenschieber einen Koppler
(32) für Eingang und Ausgang (34) und N variable Kondensatoren (10) umfasst, wobei
N ein ganzzahliger Wert von 2 oder mehr ist, wobei jeder der variablen Kondensatoren
im Gebrauch Hochfrequenzreflexion bereitstellen,
wobei jeder der variablen Kondensatoren durch mindestens einen Impedanzwandler (K,
26) mit dem Koppler verbunden ist, wobei die charakteristischen Impedanzen der Impedanzwandler
so gewählt sind, dass der Phasenschieber eine Phasenverschiebung bereitstellt, die
zumindest im wesentlichen proportional zum Wert von N ist, dadurch gekennzeichnet, dass
jeder der variablen Kondensatoren elektrochromes Material umfasst.
2. Funkfrequenzreflexionstyp-Phasenschieber nach Anspruch 1, wobei jeder der variablen
Kondensatoren (10) ein Elektrolytelement (16) und mindestens ein elektrochromes Element
(14, 18) zwischen einer ersten Elektrode (12) und einer zweiten Elektrode (20) umfasst.
3. Funkfrequenzreflexionstyp-Phasenschieber nach Anspruch 2, wobei die erste Elektrode
(12) eine Grundplatte umfasst, auf der das erste elektrochrome Element (14) liegt,
und das Elektrolytelement (16) auf dem elektrochromen Element (14) liegt, wobei das
elektrochrome Element eine elektrochrome Schicht umfasst und das Elektrolytelement
eine Elektrolytschicht umfasst.
4. Funkfrequenzreflexionstyp-Phasenschieber nach Anspruch 2 oder Anspruch 3, wobei jeder
der variablen Kondensatoren ferner ein zweites elektrochromes Element (18) zwischen
dem Elektrolytelement und der zweiten Elektrode umfasst, wobei das zweite elektrochrome
Element eine zweite elektrochrome Schicht umfasst.
5. Funkfrequenzreflexionstyp-Phasenschieber nach einem der vorhergehenden Ansprüche,
wobei der Koppler (32) ein 3dB-Koppler mit vier Anschlüssen ist, wobei N'/2 der variablen
Kondensatoren über einen von zwei der Anschlüsse mit dem Koppler verbunden sind und
N'/2 der Kondensatoren über einen zweiten der zwei Anschlüsse mit dem Koppler verbunden
sind, wobei N' eine gerade ganze Zahl von 4 oder mehr ist.
6. Funkfrequenzreflexionstyp-Phasenschieber nach einem der Ansprüche 1 bis 4, wobei der
Koppler ein Zirkulator mit drei Anschlüssen ist, wobei die N variablen Kondensatoren
über einen der Anschlüsse mit dem Zirkulator verbunden sind.
7. Funkfrequenzreflexionstyp-Phasenschieber nach einem der vorhergehenden Ansprüche,
wobei die Impedanzwandler (K, 26) Mikrostreifenleitungen sind.
8. Funkfrequenzreflexionstyp-Phasenschieber nach einem der vorhergehenden Ansprüche,
wobei die charakteristischen Impedanzen der Impedanzwandler entsprechend einem ausgewählten
Wert eines Parameterwerts
q ausgewählt sind, der für einen gegebenen Kondensator gemäß
bestimmt wird, wobei
Z0 die charakteristische Impedanz der Impedanzwandler,
Xmin die minimale Reaktanz des Kondensators und
Xmax die maximale Reaktanz des Kondensators ist.
9. Funkfrequenzreflexionstyp-Phasenschieber nach einem der vorhergehenden Ansprüche,
wobei die Kapazität jedes der variablen Kondensatoren (10) durch Einstellen einer
an die Kondensatoren angelegten Gleichspannung variabel ist.
10. Funkfrequenzreflexionstyp-Phasenschieber nach Anspruch 9, wobei die Phasenverschiebung
zumindest im Wesentlichen proportional zum Wert von N ist, wenn ein Mittelwert der
Kapazität der variablen Kondensatoren (10) so gewählt wird, dass die entsprechende
Reaktanz bei einer Betriebsfunkfrequenz die charakteristische Impedanz Z0 ist.
11. Funkfrequenzreflexionstyp-Phasenschieber nach Anspruch 10, wobei die Kondensatoren
zwischen einem "voll eingeschalteten" Zustand mit höherer Kapazität, wenn sich die
Gleichspannung auf einem ersten Pegel befindet, und einem "ausgeschalteten" Zustand
mit niedrigerer Kapazität, wenn sich die Gleichspannung auf einem zweiten Pegel befindet,
variabel sind.
12. Verfahren zur Funkfrequenzreflexionstyp-Phasenverschiebung, durch Folgendes:
Anlegen eines Eingangssignals an einen Phasenschieber (30), der einen Koppler (32)
für Eingang und Ausgang und N variable Kondensatoren (10) umfasst, wobei N ein ganzzahliger
Wert von 2 oder mehr ist, wobei jeder der variablen Kondensatoren Funkfrequenzreflexion
bereitstellt, wobei jeder der variablen Kondensatoren durch mindestens einen Impedanzwandler
mit dem Koppler verbunden ist, wobei die charakteristischen Impedanzen der Impedanzwandler
(K, 26) so gewählt sind, dass der Phasenschieber eine Phasenverschiebung bereitstellt,
die zumindest im Wesentlichen proportional zum Wert von N ist, wobei jeder der variablen
Kondensatoren elektrochromes Material umfasst; und
Empfangen eines Ausgangssignals vom Koppler.
13. Verfahren zur Funkfrequenzreflexionstyp-Phasenverschiebung nach Anspruch 12, wobei
die Kapazität jedes der variablen Kondensatoren (10) durch Einstellen einer an die
Kondensatoren angelegten Gleichspannung variabel ist.
14. Verfahren zur Funkfrequenzreflexionstyp-Phasenverschiebung nach Anspruch 13, wobei
die Phasenverschiebung zumindest im Wesentlichen proportional zum Wert von N ist,
wenn ein Mittelwert der Kapazität der variablen Kondensatoren (10) so gewählt wird,
dass die entsprechende Reaktanz bei einer Betriebsfunkfrequenz die charakteristische
Impedanz Z0 ist.
1. Déphaseur de Fréquence Radio du type à réflexion (30),
le déphaseur comprenant un coupleur (32) d'entrée et de sortie (34), et N condensateurs
variables (10), où N est un nombre entier égal à 2 ou plus, chacun des condensateurs
variables assurant, lors de l'utilisation, une réflexion de fréquence radio,
chacun des condensateurs variables étant connecté au coupleur par au moins un transformateur
d'impédance (K, 26), les impédances caractéristiques des transformateurs d'impédance
ayant été sélectionnées de manière à ce que le déphaseur assure un déphasage au moins
sensiblement proportionnel à la valeur de N, caractérisé en ce que chacun des condensateurs variables comprend un matériau électrochrome.
2. Déphaseur de fréquence radio du type à réflexion selon la revendication 1, dans lequel
chacun des condensateurs variables (10) comprend un élément d'électrolyte (16) et
au moins un élément électrochrome (14, 18) entre une première électrode (12) et une
seconde électrode (20).
3. Déphaseur de fréquence radio du type à réflexion selon la revendication 2, dans lequel
la première électrode (12) comprend une plaque de masse sur laquelle repose le premier
élément électrochrome (14), et l'élément d'électrolyte (16) repose sur l'élément électrochrome
(14), l'élément électrochrome comprenant une couche électrochrome, et l'élément d'électrolyte
comprenant une couche d'électrolyte.
4. Déphaseur de fréquence radio du type à réflexion selon la revendication 2 ou la revendication
3, dans lequel chacun des condensateurs variables comprend en outre un second élément
électrochrome (18) entre l'élément d'électrolyte et la seconde électrode, le second
élément électrochrome comprenant une seconde couche électrochrome.
5. Déphaseur de fréquence radio du type à réflexion selon l'une quelconque des revendications
précédentes, dans lequel le coupleur (32) est un coupleur à 3 dB comportant quatre
ports, N'/2 des condensateurs variables étant connectés au coupleur par l'intermédiaire
de l'un de deux des ports, et N'/2 des condensateurs étant connectés au coupleur par
l'intermédiaire d'un second desdits deux ports, où N' est un nombre entier pair égal
à 4 ou plus.
6. Déphaseur de fréquence radio du type à réflexion selon l'une quelconque des revendications
1 à 4, dans lequel le coupleur est un circulateur à trois ports, les N condensateurs
variables étant connectés au circulateur par l'intermédiaire de l'un des ports.
7. Déphaseur de fréquence radio du type à réflexion selon l'une quelconque des revendications
précédentes, dans lequel les transformateurs d'impédance (K, 26) sont des lignes à
microrubans.
8. Déphaseur de fréquence radio du type à réflexion selon l'une quelconque des revendications
précédentes, dans lequel les impédances caractéristiques des transformateurs d'impédance
sont sélectionnées en fonction d'une valeur sélectionnée d'une valeur de paramètre
q déterminée pour un condensateur donné sous la forme
où Z
0 est l'impédance caractéristique des transformateurs d'impédance, Xmin est la réactance
minimale du condensateur et Xmax est la réactance maximale du condensateur.
9. Déphaseur de fréquence radio du type à réflexion selon l'une quelconque des revendications
précédentes, dans lequel la capacité de chacun des condensateurs variables (10) peut
être amenée à varier par ajustement d'une tension continue appliquée aux bornes des
condensateurs.
10. Déphaseur de fréquence radio du type à réflexion selon la revendication 9, dans lequel
ledit déphasage est au moins sensiblement proportionnel à la valeur de N lorsqu'une
valeur moyenne de la capacité des condensateurs variables (10) est sélectionnée de
manière à ce que la réactance correspondante, à une fréquence radio de fonctionnement,
soit l'impédance caractéristique Z0.
11. Déphaseur de fréquence radio du type à réflexion selon la revendication 10, dans lequel
les condensateurs peuvent être amenés à varier entre un état "entièrement activé"
de capacité supérieure lorsque la tension continue est à un premier niveau et un état
"désactivé" de capacité inférieure lorsque la tension continue est à un second niveau.
12. Procédé de déphasage de Fréquence Radio du type à réflexion, comprenant :
l'application d'un signal d'entrée à un déphaseur (30) comprenant un coupleur (32)
d'entrée et de sortie, et N condensateurs variables (10), où N est une valeur entière
égale à 2 ou plus, chacun des condensateurs variables assurant une réflexion de fréquence
radio, chacun des condensateurs variables étant connecté au coupleur par au moins
un transformateur d'impédance, les impédances caractéristiques des transformateurs
d'impédance (K, 26) ayant été sélectionnées de manière à ce que le déphaseur assure
un déphasage au moins sensiblement proportionnel à la valeur de N, dans lequel chacun
des condensateurs variables comprend un matériau électrochrome ; et
la réception d'un signal de sortie du coupleur.
13. Procédé de déphasage de fréquence radio du type à réflexion selon la revendication
12, dans lequel la capacité de chacun des condensateurs variables (10) peut être amenée
à varier par ajustement d'une tension continue appliquée aux bornes des condensateurs.
14. Procédé de déphasage de fréquence radio du type à réflexion selon la revendication
13, dans lequel ledit déphasage est au moins sensiblement proportionnel à la valeur
de N lorsqu'une valeur moyenne de la capacité des condensateurs variables (10) est
sélectionnée de manière à ce que la réactance correspondante, à une fréquence radio
de fonctionnement, soit l'impédance caractéristique Z0.