Reactive snubber for inductive load clamp diodes

Description

Background of the Invention

[0001] The subject invention generally relates to power switching circuits utilizing semiconductors with inductive loads and, more particularly, to improvements in snubbing techniques for power diodes used to clamp inductive voltages.

[0002] In applications such as, for example, a switching power supply, a switch regulates current through an inductive load by opening and closing at a controlled rate. The switch may be mechanical or solid-state, such as a transistor or thyristor. During the time period when the switch is nonconductive, current continues to flow through the inductive load by virtue of a current path normally established by a free-wheeling or clamp diode connected across the load. The diode provides a current path and serves to limit or clamp the inductive voltage developed by the load.

[0003] When the switch is gated into conduction, with load current in the clamp diode, current is transferred from the diode to the switch at a rate which is limited only by the source voltage and the circuit inductance until the peak reverse current of the diode is reached. At this point, the diode tends to snap off at even higher rates of change of current, generating high voltage transients from the energy stored in the stray inductance of the free-wheeling current path. With fast turn-off diodes, the voltage transients are often oscillatory due to the stray inductance ringing with the diode capacitance. In power circuits, without snubbing, these transients can radiate considerable electromagnetic interference (EMI) and can also exceed the reverse blocking voltage of the diode causing self-destruction.

[0004] The recovery characteristics of a diode are depicted by the solid line on the graph of FIG. 1. The rate of change of current during commutation is a function of circuit parameters and, more specifically, of the applied voltage and the total series inductance according to the following formula:

where

is rate of change of current, E represents the magnitude of applied voltage and L represents total series inductance.

[0005] Before complete turn-off is attained, the diode current must reverse in order to sweep out the stored charge in the device. The peak reverse recovery current (Irr) is directly proportional to the rate of change of current since the area under the negative portion of the graph remains nearly constant for a given charge. In high speed switching circuits, the peak reverse currents can become excessively high and are generally snubbed by various circuit modifications.

[0006] One technique for reducing these currents is to reduce the rate of change of current as the current level approaches zero. This can be achieved by connecting a saturable reactor in series with the diode. The reactor comes out of magnetic saturation at low current levels, thereby inserting a larger series inductance, which results in the modification of the current waveform to that illustrated by the dotted line in Fig. 1. The schematic diagram of Fig. 2 is a prior art switching circuit utilizing this method. A variety of devices can be used to perform the switching function of switch 12, but commonly a bipolar power transistor would be used. The switch 12 is connected in a series circuit between a reactive load 11 and a power source 18. A control circuit (not shown) supplies control signals to regulate the on and off time intervals of switch 12 in a manner well known in the art. When switch 12 is closed, current flows from the power source 18 through the series connection of the reactive load 11 and the controlling switch 12. When switch 12 opens, the energy stored in the reactive portion of the load slowly discharges through a clamp diode 13 and a saturable reactor 14 which are serially connected across the load 11. When switch 12 recloses, load current is then commutated from the clamp diode 13 to the switch 12. In addition to assuming the load current, switch 12 also passes the reverse recovery current required to turn off clamp diode 13. Just prior to completion of this commutation, reactor 14 comes out of saturation, inserting more inductance in series with the clamp diode 13 which lowers the peak recovery currents required to turn off the diode 13. This feature allows a lower current rating for switch 12.

[0007] A snubber circuit consisting of the parallel connected combination of a diode 15 and a capacitor 16 and a series connected resistor 17 is shown connected across the switch 12 and is used to absorb some of the power losses that occur each time switch 12 turns off. The present invention proposes to modify the shunt snubber circuit, for reasons presently explained.

[0008] The major disadvantage of this technique, i.e., utilizing a simple saturating reactor, is that reactor 14 is not saturated at the moment that switch 12 turns off and, therefore, presents a high inductive impedance in the clamp diode current path. To effect a commutation of load current from switch 12 to diode 13, the voltage at the junction of switch 12 and load 11 must increase to a value which is sufficient to overcome the inductance of the unsaturated reactor 14. This voltage must exceed that of the power source in proportion to the inductance presented by reactor 14 which requires an increased voltage rating for switch 12.

Summary of the Invention

[0009] It is an object of this invention to provide an improved snubbing technique for a power clamp diode when used in a switching circuit.

[0010] It is another object of this invention to improve the turn-on speed of a power clamp diode when a saturable reactor device is used to improve the turn-off characteristics of the diode.

[0011] It is a further object of the invention to provide a temporal source of energy at the appropriate time with sufficient magnitude to improve the turn-on speed of a power diode connected in series with a saturable reactor.

[0012] The foregoing objects of the invention are achieved by modification of the conventional shunt snubber circuit such as that illustrated in Fig. 1 by providing a secondary saturation control input to the saturable reactor as is set out in claims 1 and 4. The control input is provided through a secondary winding on the reactor core. In a typical thus modified snubber circuit, the current pulse derived from the conventional part - of the shunt snubber circuit (provided for switch protection), is coupled to the secondary winding. The pulse of current in the shunt snubber occurs at the correct time and magnitude to perform the desired function. The current pulse through the secondary winding saturates the reactor core such that a low impedance is presented in the clamp diode circuit at the start of the clamp period.

Brief Description of the Drawings

[0013] The specific nature of the invention, as well as other objects, aspects, uses, and advantages thereof, will clearly appear from the following description and from the accompanying drawings, in which:

FIG. 1 is a graph of a current waveform during the recovery time of a typical power diode;

FIG. 2 is a schematic diagram of a prior art power switching circuit with a conventional saturable reactor;

FIG. 3 is a schematic diagram of a power switching circuit according to the present invention; and

FIG. 4 is a simplified schematic diagram illustrating one embodiment of the present invention as applied to a half-bridge power inverter switching circuit configuration.

Detailed Description

[0014] Reference is now made to FIG. 3 of the drawings wherein like reference numerals represent the same or similar elements as shown in the prior art circuit diagram of FIG. 2. The snubber circuit has been modified by the addition of a secondary winding 14A on the core of reactor 14, which winding 14A is connected in series with the diode 15. In typical power switching applications, such as choppers and inverters, the switching frequency is usually high enough to maintain continuous current flowing in the load 11 and in some applications may be several thousand cycles per second. When switch 12 is on, current flows through the source 18, load 11, and switch 12 and increases toward some maximum value determined by the magnitude of source voltage and load impedance. As the switch 12 opens, the load current diverts around switch 12 through the secondary winding 14A of reactor 14, diode 15, and capacitor 16 while an insignificant portion flows through resistor 17. This path for the load current is maintained until capacitor 16 becomes charged to slightly above the magnitude of the voltage of power source 18. At this point, diode 13 becomes forward biased and conducts current rapidly, transferring the load current from the snubber path to the clamp diode 13 path. Since the two windings on reactor 14, i.e., the primary winding 14B and secondary winding 14A, represent a transformer when the core is not in saturation, the polarity sense of the windings, as represented by the dots on the schematic diagram, must be observed. With the polarity sense as shown by the dots, the voltage induced across the primary winding 14B due to the initial increasing load current in the secondary winding 14A is of the polarity to reverse bias diode 13, keeping it off. The induced voltage on the primary winding 14B disappears as soon as the load current flowing in the secondary winding 14A is of sufficient magnitude to saturate the core of reactor 14. Since the reactor 14 is saturated before the voltage at the junction of load 11 and switch 12 exceeds the power source 18 voltage, diode 13 starts conducting current immediately as it becomes forward biased. When diode 13 conducts, the load current rapidly transfers from the secondary winding 14A to the primary winding 14B and the flux density, due to the net ampere- turns, increases further into saturation by an amount proportional to the turns ratio.

[0015] The current from load 11 flowing through the primary winding 14B maintains reactor 14 in saturation until switch 12 turns on again. When switch 12 conducts, the load current rapidly transfers from the clamp diode 13 path to the switch 12 path. As the current in the primary winding 14B approaches zero, the core of reactor 14 comes out of saturation which increases the inductance of the primary winding 14B slowing the rate of change of commutating current as in a simple conventional reactor. The voltage which builds up across the primary winding 14B as reactor 14 comes out of saturation is transformed to the secondary winding 14A. This voltage is made insufficient to overcome the reverse bias on the snubber diode 15 by appropriately selecting the turns ratio between windings 14B and 14A to step down the induced voltage. The reverse bias on diode 15 supplied by the charge on capacitor 16 is lowered when switch 12 is turned on by discharging capacitor 16 through resistor 17 and switch 12. This discharge time constant is made much slower than the commutation time of load current from clamp diode 13 to switch 12, thereby maintaining a significant reverse bias on diode 15 during the commutation.

[0016] The parameters for the saturable reactor are selected to minimize the inductance in series with the snubber diode 15 and to allow saturation by the load current flowing in the snubber path prior to commutation. The number of primary turns is selected to provide sufficient snubbing of the clamp diode 13 recovery characteristics when the core of reactor 14 is not in saturation. The resistor 17 is provided only to permit discharge of capacitor 16 and could be connected across diode 15 alone.

[0017] A specific embodiment of the invention as it applies to half-bridge inverter configuration is shown in FIG. 4, wherein like reference numerals perform the same or similar function of FIG. 3. Each switching device 12a and 12b is supplied with on/off switching command signals on alternate half cycles of a base frequency. All the components with an "a" designation perform the functions described for FIG. 3 during one half cycle of the base frequency, while the components with a "b" designation perform these functions on the alternate half cycles of the base frequency. Since each half of the half-bridge inverter of FIG. 4 operates independently in the manner described with respect to FIG. 3, no additional explanation of the circuit of FIG. 4 is believed necessary. However, it should be noted that when switch 12a opens, the load current circulates through power source 18b as is typical in prior art half-bridge inverters. The switching frequency for each switch 12 is typically several multiples of the base frequency and the switch 12 may be operated in a pulse width modulation mode to produce the desired base frequency response at load 11.

[0018] It will be recognized by those skilled in the art that FIG. 4 represents a half-bridge configuration, and that it is possible to provide a full-bridge configuration by simply replacing the power source center tap by two more identical power switches. Likewise, it is possible to extend this basic circuit to multi-phase systems by merely adding additional switching devices 12 arranged to connect each additional power phase to the load 11 at appropriate times.

Claims

1. A snubber circuit for use in a power switching circuit including at least one power switch and power clamp diode (13), the switching circuit being of the type responsive to control signals for connecting and disconnecting an inductive load (11) from a power source (18) comprising:

a) a saturable reactor (14) having a primary winding (14b) and a secondary winding (14a), said primary winding (14b) being connected in series with the clamp diode (13) across the load (11), the clamp diode (13) being poled so as to permit continuous load current to flow when the load (11) is disconnected from the power source (18) by the power switch (12); and

b) a shunt snubber circuit connected across the power switch (12), the shunt snubber circuit serving to supply a current pulse to said secondary winding (14a) in a manner to force rapid magnetic saturation of said saturable reactor (14) when the load (11) is disconnected from the power source (18) by the power switch (12) whereby load current is rapidly transferred into the clamp diode (13) immediately upon the clamp diode (13) being forward biased.

2. The snubber circuit of Claim 1 wherein said shunt snubber circuit connected across said power switch (12) comprises:

a) a diode (15);

b) a capacitor (16);

c) means for connecting said diode (15), said capacitor (16) and said secondary winding (14a) in a series current path in parallel with the power switch (12), said diode (15) being poled to provide a current path for continuous current through the load (11);

d) a resistor (17); and

e) means for connecting said resistor (17) in parallel circuit with the series combination of said secondary winding (14a) and said diode (15) to thereby provide a discharge current path for said capacitor (16) when the switch (12) is conductive, whereby termination of conduction of the power switch (12) causes a current pulse to flow through said path formed by said secondary winding (14a), said diode (15) and said capacitor (16) to force saturation of said reactor (14).

3. The snubber circuit of Claim 2 wherein the power switch (12) comprises a bipolar transistor.

4. A snubber circuit for use in a switching power regulator arranged in a half-bridge inverter configuration for regulating current in an inductive load (11), the regulator including first and second series connected switching devices (12a, 12b) and means for rendering the devices alternately conductive, first and second series connected power sources (18a, 18b) coupled across the respective switching devices, the inductive load (11) being connected between a junction intermediate the power sources (18a, 18b) and a junction intermediate the switching devices (12a, 12b), the snubber circuit comprising:

a) first and second saturable reactors (19a, 19b), each having a primary winding and a secondary winding;

b) first and second clamping diodes (13a, 13b);

c) means for connecting said primary winding of said first reactor (19a) and said first diode (13a) in a series circuit across the second switching device (12b);

d) means for connecting said primary winding of said second reactor (19b) and said second diode (13b) in a series circuit across the first switching device (12a);

e) means for supplying a current pulse to said secondary winding of said first reactor (19a) concurrent with opening of the first switching device (12a) to thereby force rapid saturation of said first reactor (19a) to enable rapid transfer of current to a path through said first clamp diode (13a); and

f) means for supplying a current pulse to said secondary winding of said second reactor (19b) concurrent with opening of the second switching device (12b) to thereby force rapid saturation of said second reactor (19b) to enable rapid transfer of current to a path through said second clamp diode (13b) (Fig. 4).

5. The snubber circuit of Claim 4 wherein each of said means for supplying a current pulse comprises:

a) a resistor (17a, 17b);

b) a capacitor (16a, 16b);

c) a diode (15a, 15b);

d) means connecting said resistor (17a, 17b) and said capacitor (16a, 16b) in a series circuit across a corresponding one of said switching devices (12a, 12b); and

e) means connecting said diode (15a, 15b) and a corresponding one of said secondary windings in a series current path across said resistor (17a, 17b), whereby termination of conduction of said corresponding one of said switching devices (12a, 12b) generates a current pulse through said path formed by said diode (15a, 15b), said secondary winding and said capacitor (16a, 16b) (Fig. 4).

6. The snubber circuit of Claim 5 wherein the switching device (12a, 12b) comprises a bipolar transistor.

Revendications

1. Circuit amortisseur destiné à être utilisé dans un circuit de commutation de puissance comprenant au moins un élément de commutation de puissance et une diode de limitation de puissance (13), le circuit de commutation étant du type qui réagit à des signaux de commande en connectant une charge inductive (11) à une source d'alimentation (18) et en déconnectant cette charge de la source, caractérisé en ce qu'il comprend: (a) une réactance saturable (14) ayant un enroulement primaire (14b) et un enroulement secondaire (14a), l'enroulement primaire (14b) étant connecté en série avec la diode de limitation (13) aux bornes de la charge (11), et la diode de limitation (13) étant orientée de façon à permettre la circulation continue du courant de la charge (11) lorsque cette dernière est déconnectée de la source d'alimentation (18) par l'élément de commutation de puissance (12); et (b) un circuit amortisseur shunt connecté aux bornes de l'élément de commutation de puissance (12), ce circuit amortisseur shunt appliquant une impulsion de courant à l'enroulement secondaire (14a) de manière à forcer une saturation magnétique rapide de la réactance saturable (14) lorsque la charge (11) est déconnectée de la source d'alimentation (18) par l'élément de commutation de puissance (12), grâce à quoi le courant de la charge est rapidement transféré vers la diode de limitation (13), dès que la diode de limitation (13) est polarisée en sens direct.

2. Circuit amortisseur selon la revendication 1, caractérisé en ce que le circuit amortisseur shunt connecté aux bornes de l'élément de commutation de puissance (12) comprend: (a) une diode (15); (b) un condensateur (16); (c) des moyens destinés à connecter la diode (15), le condensateur (16) et l'enroulement secondaire (14a) en un circuit série, en parallèle sur l'élément de commutation de puissance (12), la diode (15) étant orientée de façon à établir un circuit assurant la circulation permanente d'un courant dans la charge (11); (d) un résistance (17); et (e) des moyens destinés à connecter cette résistance (17) en parallèle avec la combinaison série de l'enroulement secondaire (14a) et de la diode (15), pour établir ainsi un circuit de décharge pour le condensateur (16) lorsque l'élément de commutation (12) est conducteur, grâce à quoi la terminaison de la conduction de l'élément de commutation de puissance (12) fait circuler une impulsion de courant dans le circuit formé par l'enroulement secondaire (14a), la diode (15) et le condensateur (16), de façon à forcer la saturation de la réactance (14).

3. Circuit amortisseur selon la revendication 2, caractérisé en ce que l'élément de commutation de puissance (12) consiste en un transistor bipolaire.

4. Circuit amortisseur destiné à l'utilisation dans un régulateur de puissance à commutation ayant une configuration d'onduleur en demi- point, prévu pour réguler le courant dans une charge inductive (11), le régulateur comprenant des premier et second dispositifs de commutation (12a, 12b) connectés en série, et des moyens pour faire passer alternativement ces dispositifs à l'état conducteur, des première et seconde sources d'alimentation (18a, 18b) connectées en série et branchées aux bornes des dispositifs de commutation respectifs, la charge inductive (11) étant connectée entre un point de connexion situé entre les sources d'alimentation (18a, 18b), et un point de connexion situé entre les dispositifs de commutation (12a, 12b), caractérisé en ce qu'il comprend: (a) des première et seconde réactances saturables (19a, 19b), ayant chacune un enroulement primaire et un enroulement secondaire; (b) des première et seconde diodes de limitation (13a, 13b); (c) des moyens destinés à connecter l'enroulement primaire de la première réactance (19a) et la première diode (13a) en un circuit série aux bornes du second dispositif de commutation (12b); (d) des moyens destinés à connecter l'enroulement primaire de la seconde réactance (19b) et la seconde diode (13b) en un circuit série aux bornes du premier dispositif de commutation (12a); (e) des moyens destinés à appliquer une impulsion de courant à l'enroulement secondaire de la première réactance (19a), simultanément au blocage du premier dispositif de commutation (12a), pour forcer ainsi la saturation rapide de la première réactance (19a), afin de permettre un transfert de courant rapide vers un chemin passant par la première diode de limitation (13a); et (f) des moyens destinés à appliquer un impulsion de courant à l'enroulement secondaire de la seconde réactance (19b), simultanément au blocage du second dispositif de commutation (12b), pour forcer ainsi la saturation rapide de la seconde réactance (19b), afin de permettre un transfert de courant rapide vers un chemin passant par la seconde diode de limitation (13b) (figure 4).

5. Circuit amortisseur selon la revendication 4, caractérisé en ce que chacun des moyens destinés à appliquer un impulsion de courant comprend: (a) une résistance (17a, 17b); (b) un condensateur (16a, 16b); (c) une diode (15a, 15b); (d) des moyens connectant cette résistance (17a, 17b) et ce condensateur (16a, 16b) en un circuit série branché aux bornes de l'un correspondant des dispositifs de commutation (12a, 12b); et (e) des moyens connectant cette diode (15a, 15b) et l'un correspondant des enroulements secondaires en un circuit série branché aux bornes de la résistance (17a, 17b), grâce à quoi la terminaison de la conduction du dispositif de commutation correspondant (12a, 12b) génère une impulsion de courant dans le circuit formé par la diode (15a, 15b), l'enroulement secondaire et le condensateur (16a, 16b) (figure 4).

6. Circuit amortisseur selon la revendication 5, caractérisé en ce que le dispositif de commutation (12a, 12b) consiste en un transistor bipolaire.

Ansprüche

1. Dämpfungsschaltung zur Verwendung in einem Leistungsschaltkreis, der wenigstens einen Leistungsschalter und eine Leistungsklemmdiode (13) aufweist und der auf Steuersignale anspricht zum Verbinden und Trennen einer induktiven Last (11) mit bzw. von einer Leistungsquelle (18), enthaltend:

. a) eine sättigbare Drossel (14) mit einer Primärwicklung (14b) und einer Sekundärwicklung (14a), wobei die Primärwicklung (14b) mit der Klemmdiode (13) in Reihe der Last (11) parallel geschaltet ist und die Klemmdiode (13) derart gepolt ist, daß ein kontinuierlicher Laststrom fließen kann, wenn die Last (11) durch den Leistungsschalter (12) von der Leistungsquelle (18) getrennt ist, und

b) eine Paralleldämpfungsschaltung, die dem Leistungsschalter (12) parallel geschaltet ist und dazu dient, der Sekundärwicklung (14a) einen Stromimpuls in der Weise zuzuführen, daß eine schnelle magnetische Sättigung der sättigbaren Drossel (14) erzwungen wird, wenn die Last (11) durch den Leistungsschalter (12) von der Leistungsquelle (18) getrennt ist, wodurch Laststrom schnell auf die Klemmdiode (13) übertragen wird unmittelbar nach-dem die Klemmdiode (13) in Durchlaßrichtung vorgespannt ist.

2. Dämpfungsschaltung nach Anspruch 1, wobei die dem Leistungsschalter (12) parallel geschaltete Paralleldämpfungsschaltung aufweist:

a) eine Diode (15),

b) einen Kondensator (16),

c) Mittel zum Verbinden der Diode (15), des Kondensators (16) und der Sekundärwicklung (14a) in einem Reihenstrompfad parallel zu dem Leistungsschalter (12), wobei die Diode (15) so gepolt ist, daß ein Strompfad für einen kontinuierlichen Strom durch die Last (11) gebildet ist,

d) einen Widerstand (17) und

e) Mittel zum Verbinden des Widerstands (17) in einem Parallelkreis mit der Reihenschaltung aus der Sekundärwicklung (14a) und der Diode (15), um dadurch einen Entladestrompfad für den Kondensator (16) zu bilden, wenn der Schalter (12) leitend ist, wodurch eine Beendigung der Leitfähigkeit des Leistungsschalter (12) bewirkt, daß ein Stromimpuls durch den Pfad, der von der Sekundärwicklung (14a), der Diode (15) und dem Kondensator (16) gebildet ist, fließt, um eine Sättigung der Drossel (14) zu bewirken.

3. Dämpfungsschaltung nach Anspruch 2, dadurch gekennzeichnet, daß der Leistungsschalter

(12) einen bipolaren Transistor aufweist.

4. Dämpfungsschaltung zur Verwendung in einem Leistungsschaltregler, der in einer Halbbrücken-Wechselrichterkonfiguration angeordnet, ist, zum Regeln des Stromflusses in einer induktiven Last (11), wobei der Regler erste und zweite in Reihe geschaltete Schaltvorrichtungen (12a, 12b) und Mittel zum abwechselnden Durchschalten der Vorrichtungen und erste und zweite in Reihe geschaltete Leistungsquellen (18a, 18b) aufweist, die den entsprechenden Schaltvorrichtungen parallel geschaltet sind, wobei die induktive Last (11) zwischen einen Knotenpunkt zwischen den Leistungsquellen (18a, 18b) und einen Knotenpunkt zwischen den Schaltvorrichtungen (12a, 12b) geschaltet ist, wobei die Dämpfungsschaltung enthält:

a) erste und zweite sättigbare Drosseln (19a, 19b), die jeweils eine Primärwicklung und eine Sekundärwicklung aufweisen,

b) erste und zweite Klemmdioden (13a, 13b),

c) Mittel zum Verbinden der Primärwicklung der ersten Drossel (19a) und der ersten Diode (13a) in einer Reihenschaltung parallel zu der zweiten Schaltvorrichtung (12b),

d) Mittel zum Verbinden der Primärwicklung der zweiten Drossel (19b) und der zweiten Diode (13b) in einer Reihenschaltung parallel zur ersten Schaltvorrichtung (12a),

e) Mittel zum Zuführen eines Stromsimpulses zu der zweiten Sekundärwicklung der ersten Drossel (19a) gleichzeitig mit dem Öffnen der ersten Schaltvorrichtung (12a), um dadurch eine schnelle Sättigung der ersten Drossel (19a) zu erzwingen, um einen schnellen Übergang des Stromflusses auf einen Pfad durch die erste Klemmdiode (13a) zu ermöglichen, und

f) Mittel zum Zuführen eines Stromimpulses zu der Sekundärwicklung der zweiten Drossel (19a) gleichzeitig mit dem Öffnen der zweiten Schaltvorrichtung (12b), um dadurch eine schnelle Sättigung der zweiten Drossel (19b) zu erzwingen, um einen schnellen Übergang des Stromes auf einem Pfad durch die zweite Klemmdiode (13b) zu ermöglichen (Fig. 4).

5. Dämpfungsschaltung nach Anspruch 4, wobei jedes Mittel zum Zuführen eines Stromimpulses aufweist:

a) einen widerstand (17a, 17b),

b) einen Kondensator (16a, 16b),

c) eine Diode (15a, 15b),

d) Mittel zum Verbinden des Widerstands (17a, 17b) und des Kondensators (16a, 16b) in einer Reihenschaltung parallel zu einer entsprechenden der Schaltvorrichtungen (12a, 12b) und

e) Mittel zum Verbinden der Diode (15a, 15b) und einer entsprechenden Sekundärwicklung in einer Reihenstrombahn parallel zum Widerstand (17a, 17b), wodurch eine Beendigung der Leitfähigkeit einer entsprechenden Schaltvorrichtung (12a, 12b) einen Stromimpuls durch die Bahn erzeugt, die durch die Diode (15a, 15b), die Sekundärwicklung und den Kondensator (16a, 16b) gebildet ist (Fig. 4).

6. Dämpfungsschaltung nach Anspruch 5, wobei die Schaltvorrichtung (12a, 12b) einen bipolaren Transistor aufweist.

Drawing