The present invention relates generally to data communication, and more specifically to techniques for performing beam-steering and beam-forming for wideband MIMO/MISO systems.

A multiple-input multiple-output (MIMO) communication system employs multiple (N_T) transmit antennas and multiple (N_R) receive antennas for data transmission. A MIMO channel formed by the N_T transmit and N_R receive antennas may be decomposed into N_S independent channels, with N_S ≤ min {N_T, N_R}. Each of the N_S independent channels is also referred to as a spatial subchannel or eigenmode of the MIMO channel.

A multiple-input single-output (MISO) communication system employs multiple (N_T) transmit antennas and a single receive antenna for data transmission. A MISO channel formed by the N_T transmit and single receive antenna includes a single spatial subchannel or eigenmode. However, the multiple transmit antennas may be used to provide transmit diversity or to perform beam-forming or beam-steenng for the data transmission.

For a wideband system, orthogonal frequency division multiplexing (OFDM) may be used to effectively partition the overall system bandwidth into a number of (N_F) orthogonal subbands, which are also referred to as frequency bins or subchannels. With OFDM, each subband is associated with a respective subcarrier upon which data may be modulated. For a MIMO/MISO system that utilizes OFDM (i.e., a MIMO/MISO-OFDM system), each subband of each spatial subchannel may be viewed as an independent transmission channel.

The spatial subchannel(s) of a wideband MIMO/MISO system may encounter different channel conditions due to various factors such as fading and multipath. Each spatial subchannel may experience frequency selective fading, which is characterized by different channel gains at different frequencies of the overall system bandwidth. This may then result in different signal-to-noise ratios (SNRs) at different frequencies of each spatial subchannel. Moreover, the channel conditions may deteriorate to a level where most of the spatial subchannels are highly degraded. In these situations, improved performance may be achieved by using only the best spatial subchannel for data transmission.

International Patent Publication No. WO 02/05506 discloses multiplexing of real time services and non-real time services for OFDM systems.

US Patent No. 6,377,631 discloses a transmitter incorporating spatio-temporal processing.

WO 03/073646 (published on 4-09-2003 and therefore prior art according to Article 54(3) EPC) discloses techniques to transmit data on a number of transmission channels in a multi-channel communication system using multiple transmission schemes requiring less channel state information.

There is therefore a need in the art for techniques to process data for transmission on a single spatial subchannel when warranted by the channel conditions.

Techniques are provided herein to set out in the appended claims to transmit data on a single spatial subchannel (or eigenmode) in a wideband multiple-input system, which may be a MIMO or MISO system (e.g., a MIMO-OFDM or MISO-OFDM system). These techniques may be used to provide improved performance under adverse channel conditions.

Data transmission on a single eigenmode (typically the best or principal eigenmode for a MIMO system) may be achieved using beam-steering or beam-forming. For a wideband MIMO/MISO system, the beam-steering or beam-forming is performed for each subband that is selected for use for data transmission based on a steering vector obtained for that subband. The beam-steering or beam-forming may also be performed in conjunction with a particular power allocation scheme that allocates the total transmit power to the subbands.

In an embodiment, a method is provided to process data for transmission via a single eigenmode of a multiple-input channel (e.g.. a MIMQ or MISO channel). In accordance with the method, a steering vector is obtained for each of a number of subbands. Each steering vector includes N_T elements for N_T transmit antennas. Depending on how the steering vectors are defined, beam-steering or beam-forming can be achieved for each subband.

The total transmit power is allocated to the subbands based on a particular power allocation scheme (e.g., full channel inversion, selective channel inversion, water-filling, or uniform, all of which are described below). A scaling value is then obtained for each subband based on the transmit power allocated to the subband.

Data to be transmitted is coded and modulated based on one or more coding and modulation schemes to provide modulation symbols. The modulation symbols to be transmitted on each subband are then scaled with the subband's scaling value, and the scaled modulation symbols are further preconditioned with the subband's steering vector A stream of preconditioned symbols is then formed for each transmit antenna, and this stream is further processed to generate a modulated signal suitable for transmission from a respective transmit antenna

Various aspects and embodiments of the invention are described in further detail below The invention further provides methods, program codes, digital signal processors, transmitter unfits, receiver units, and other apparatuses and elements that implement various aspects, embodiments, and features of the invention, as described in further detail below.

The features, nature, and advantages of the present invention will become more apparent from the detailed description set forth below when taken in conjunction with the drawings in which like reference characters identify correspondingly throughout and wherein:

• PIG. 1 graphically illustrates the results of eigenvalue decomposition for a number of subbands in a MIMO-OFDM system;
• FIG. 2 is a block diagram of a transmitter system and a receiver system in the MIMO-OFDM system;
• FIG. 3 is a block diagram of a transmitter unit within the transmitter system;
• FIG. 4 is a block diagram of a signal scaling unit, a beam-steering unit, and a multiplexer within the transmitter unit; and
• FIG. 5 is a flow diagram for processing data for transmission on a single eigenmode of a multiple-input channel using beam-steenng or beam-forming.

The beam-steering and beam-forming techniques described herein may be used in various wideband MlMO/MISO communication systems. For clarity, these techniques are described specifically for a MIMO-OFDM system that effectively divides the overall system bandwidth into N_F orthogonal subbands.

The model for the MIMO-OFDM system may be expressed as: $$\underline{\mathbf{y}}\left(k\right)=\underline{\mathbf{H}}\left(k\right)\underline{\mathbf{x}}\left(k\right)+\underline{\mathbf{n}}\left(k\right),\mathrm{for}k\in \left\{1,\dots ,N_F\right\},$$
where

y(k)

is a vector with N_R entries, {y_i (k)} for i ∈ {1,..., N_R}, for the symbols received via the N_R receive antennas for the k-th subband (i.e., the "recede" vector);

x(k)

is a vector with N_T entries, {x_j(k)} for j ∈ {1,..., N_T}, for the symbols transmitted from the N_T transmit antennas for the k-th subband (i.e., the "transmit" vector);

H(k)

is an (N_R×N_T) channel response matrix with entries, {h_ij(k)} for i ∈ {1,..., N_R} and j ∈ {1,..., N_T}, which are the complex gains from the N_T transmit antennas to the N_R receive antennas for the k-th subband; and

n(k)

is additive white Gaussian noise (AWGN) for the k-th subband, with zero mean and a covariance matrix of Δ_n = σ²I, where I is the identity matrix and σ² is the noise variance.

For simplicity, each subband is assumed to be frequency non-selective (i.e., with a flat frequency response across the entire subband). In this case, the channel response h_ij(k) for each transmission channel can be represented by a single complex value, and the elements of the channel response matrix H(k) are scalars. Also for simplicity, the noise variance is assumed to be constant across all transmission channels. For a time division duplexed (TDD) system, the forward and reverse links share the same system bandwidth and each subband may be assumed to be reciprocal. That is, if H(k) represents the channel response matnx from antenna array A to antenna array B, then a reciprocal channel implies that the coupling from array B to array A is given by H^H(k).

The channel response matrix H(k) for each subband may be "diagonalized" to obtain the N_S independent channels for that subband This can be achieved by performing eigenvalue decomposition on the correlation matrix of H(k), which is R(k) = H^H (k)H(k), where H^H(k) denotes the conjugate transpose of H(k). The eigenvalue decomposition of the correlation matrix R(k) may be expressed as: $$\underline{\mathbf{R}}\left(k\right)=\underline{\mathbf{E}}\left(k\right)\underline{\mathbf{D}}\left(k\right){\underline{\mathbf{E}}}^H\left(k\right),\mathrm{for}k\in \left\{1,\dots ,N_F\right\},$$
where

E(k)

is an (N_T×N_T) unitary matrix whose columns are the eigenvectors of R(k) ; and

D(k)

is an (N_T×N_T) diagonal matrix with entries on the diagonal corresponding to the eigenvalues of R(k).

The eigenvalue decomposition may also be performed using singular value decomposition, as is known in the art.

The diagonal matrix D(k) for each subband contains non-negative real values along the diagonal and zeros everywhere else. These diagonal entries are referred to as the eigenvalues of R(k) and are related to the complex gains for the independent channels (or eigenmodes) of the MIMO channel for the k-th subband. Since the number of independent channels is N_S ≤ min {N_T, N_R} for a MIMO system with N_T transmit and N_R receive antennas, there are N_S non-zero eigenvalues of R(k). The eigenvalues of R(k) are denoted as {λ_i (k)}, for i = {1, ..., N_S} and k = {1, .., N_F}.

For the MIMO-OFDM system, the eigenvalue decomposition may be performed independently for the channel response matrix H(k) for each subband to determine the N_S eigenmodes for that subband The N_S eigenvalues for each diagonal matrix D(k), for k ∈ {1,... N_F}, may be ordered such that {λ₁(k) ≥ λ₂(k) ≥ ... ≥ λ_NS(k)}, where λ₁(k) is the largest eigenvalue and λ_NS(k) is the smallest eigenvalue for the k-th subband.

FIG. 1 graphically illustrates the results of the eigenvalue decomposition for the N_F subbands in the MIMO-OFDM system. The set of diagonal matrices, D(k) for k = {1, ..., N_F}, is shown arranged in order along an axis 110 that represents the frequency dimension. The eigenvalues, {λ_i(k)} for i = {1, ..., N_S}, of each matnx D(k) are located along the diagonal of the matrix. Axis 112 may thus be mewed as representing the spatial dimension. Eigenmode i for all subbands (or simply, eigenmode i) is associated with a set of elements, {λ_i(k)} for k = {1, ..., N_F}, which is indicative of the frequency response across the N_F subbands for that eigenmode. The set of elements {λ_i(k)} for each eigenmode is shown by the shaded boxes along a dashed line 114. Each shaded box in FIG. 1 represents a transmission channel. For each eigenmode that experiences frequency selective fading, the elements {λ_i(k)} for that eigenmode may be different for different values of k.

If the eigenvalues in each diagonal matnx D(k) are sorted in descending order, then eigenmode 1 (which is also referred to as the principal eigenmode) would include the largest eigenvalue in each matrix, and eigenmode N_S would include the smallest eigenvalue in each matrix.

Under adverse channel conditions, most of the eigenmodes may be highly degraded. In these situations, improved performance may be achieved by using only the best eigenmode (i.e., the principal eigenmode) for data transmission.

The model for a MISO-OFDM system may be expressed as: $$y\left(k\right)=\underline{\mathbf{h}}\left(k\right)\underline{\mathbf{x}}\left(k\right)+n\left(k\right),\mathrm{for}k\in \left\{1,\dots ,N_F\right\},$$
where

y(k)

denotes the symbol received on the k-th subband;

x(k)

is a vector with N_T entries for the symbols transmitted from the N_T transmit antennas for the k-th subband;

h(k)

is a (1×N_T) channel response vector with entries, {h_j(k)} for j ∈ {1, ..., N_T}, which are the complex gains from the N_T transmit antennas to the single receive antenna for the k-th subband; and

n(k)

is additive white Gaussian noise (AWGN) for the k-th subband.

For MIMO and MISO systems, data transmission on a single eigenmode may be achieved using beam-steenng or beam-forming, both of which are described below.

The beam-forming technique transmits data on a single (e.g., principal) eigenmode by preconditioning the modulation symbols with the eigenvector for this eigenmodes. For the MIMO-OFDM system, the beam-forming is performed for each subband using the eigenvectors obtained for that subband.

In equation (2), the unitary matrix E(k) contains N_T columns for N_T eigenvectors, i e., E(k) = [e₁ (k) e₂ (k) .. e_NT(k)]. The eigenvectors are also referred to as steering vectors. Each eigenvector is associated with a respective eigenmode and eigenvalue of the diagonal matrix D(k) (i.e., eigenvector e_i(k) is associated with eigenvalue λ_i(k) for subband k). When the eigenvalues of D(k) are sorted in descending order as described above, the eigenvectors of E(k) are also rearranged in the corresponding order. After the sorting/rearrangement, eigenvector e_i(k) corresponds to the largest eigenvalue λ₁(k) and is the eigenvector for the principal eigenmodes for the k-th subband. This eigenvectors e₁(k) includes N_T elements for the N_T transmit antennas and can be expressed as: $${\underline{\mathbf{e}}}_1\left(k\right)={\left[\begin{array}{cccc}{e}_{1,1}\left(k\right)& e_{1,2}\left(k\right)& \dots & e_{1,N_T}\left(k\right)\end{array}\right]}^T,\mathrm{for}k\in \left\{1,\dots ,N_F\right\},$$

• where "T" denotes the transpose.

The preconditioning at the transmitter to achieve beam-forming for each subband may be expressed as: $$\underline{\mathbf{x}}\left(k\right)=\sqrt{P\left(k\right)}{\underline{\mathbf{e}}}_1\left(k\right)s\left(k\right),\mathrm{for}k\in \left\{1,\dots ,N_F\right\},$$
where
s(k)  is the modulation symbol to be transmitted on the k-th subband;
$\sqrt{P\left(k\right)}$  is a scaling value derived based on the transmit power P(k) allocated to the k-th subband; and
x(k)  is the transmit vector with N_T preconditioned symbols for the k-th subband.

As shown in equation (4), the beam-forming technique generates one transmit vector x(k) for each subband based on the eigenvector e₁(k) for the principal eigenmode. Since the elements of the eigenvector e₁(k) may have different magnitudes, the elements of the transmit vector x(k) may also have different magnitudes.

For each transmit antenna i, the N_F preconditioned symbols to be transmitted on the N_F subbands in symbol period n are multiplexed together into a (per-antenna transmit) vector x_i(n), which may be expressed as: $${\underline{\mathbf{x}}}_i\left(n\right)={\left[\begin{array}{cccc}{e}_{1,i}\left(1\right)\tilde{s}\left(1\right)& e_{1,i}\left(2\right)\tilde{s}\left(2\right)& \dots & e_{1,i}\left(N_F\right)\tilde{s}\left(N_F\right)\end{array}\right]}^T,\mathrm{for}i\in \left\{1,\dots ,N_T\right\},$$

• where s̃(k) is a scaled modulation symbol and given as $\tilde{s}\left(k\right)=\sqrt{P\left(k\right)}s\left(k\right)\mathrm{.}$

For the MISO-OFDM system, the beam-forming is also performed for each subband using the steering vector obtained for that subband. If the channel decomposition is performed on the channel response vector h(k), the result will be one eigenmode (i.e., one non-zero value for the matrix D(k)) and one steering vector. This steering vector will be equal to h*(k). The beam-forming for MISO may be performed as shown in equation (4).

The beam-steenng technique transmits data on the principal eigenmode by preconditioning the modulation symbols with a "normalized" steering vector for this eigenmode The beam-steering is also performed for each subband for the MIMO-OFDM system.

As noted above, the elements of each eigenvectors e₁(k), for k ∈ {1, .., N_F}, for the principal eigenmode may have different magnitudes. Consequently, the per-antenna transmit vectors x_i (n), for i ∈ {1, ..., N_T}, may have different magnitudes. If the transmit power for each transmit antenna is limited (e.g., because of limitations of the power amplifiers), then the beam-forming technique may not fully use the total power available for each antenna.

The beam-steering technique uses only the phase information from the eigenvectors e₁(k), for k ∈ {1, ..., N_F}, and normalizes each transmit steering vector such that all N_T elements have equal magnitudes. The normalized steering vector ẽ(k) for the k-th subband may be expressed as: $$\underline{\tilde{\mathbf{e}}}\left(k\right)={\left[\begin{array}{cccc}A{e}^{j{\theta }_1\left(k\right)}& A{e}^{j{\theta }_2\left(k\right)}& \dots & A{e}^{j{\theta }_{N_T}\left(k\right)}\end{array}\right]}^T,$$

• where A is a constant (e.g., A = 1); and
• θ_i(k) is the phase for the k-th subband of the i-th transmit antenna, which is given as:

$${\theta }_i\left(k\right)=\angle e_{1,i}\left(k\right)={\tan }^{-1}\left(\frac{\mathrm{Im}\left\{e_{1,i}\left(k\right)\right\}}{\mathrm{Re}\left\{e_{1,i}\left(k\right)\right\}}\right)\mathrm{.}$$

The preconditioning at the transmitter to achieve beam-steering for each subband may be expressed as: $$\underline{\mathbf{x}}\left(k\right)=\sqrt{P\left(k\right)}\underline{\tilde{\mathbf{e}}}\left(k\right)s\left(k\right),\mathrm{for}k\in \left\{1,\dots ,N_F\right\}\mathrm{.}$$
As shown in equations (5a) and (5b), the elements of the normalized steering vector ẽ(k) for each subband have equal magnitude but possibly different phases. The beam-steenng technique generates one transmit vector x(k) for each subband, with the elements of x(k) having the same magnitude but possibly different phases.

As described above, for each transmit antenna i, the N_F preconditioned symbols to be transmitted on the N_F subbands in symbol period n are multiplexed together into a per-antenna transmit vector x_i(n). Since each transmits vector x_i(n), for i ∈ {1, ..., N_T}, includes the same set of scaled modulation symbols (but possibly with different phases), the total available transmit power for each antenna may be fully used.

At the receiver, to obtam an estimate of the modulation symbol s(k), the received vector y(k) for each subband may be pre-multiplied (or "conditioned") with either ẽ^H(k)H^H(k) (if beam-steering was performed) or ${\underline{\mathbf{e}}}_1^H\left(k\right){\underline{\mathbf{H}}}^H\left(k\right)$ (if beam-forming was performed). If beam-steering was performed, then the conditioning to obtain the symbol estimate ŝ(k) may be expressed as: $$\begin{array}{ll}\hat{s}\left(k\right)& ={\underline{\tilde{\mathbf{e}}}}^H\left(k\right){\underline{\mathbf{H}}}^H\left(k\right)\underline{\mathbf{y}}\left(k\right)\\ & =\sqrt{P\left(k\right)}{\underline{\tilde{\mathbf{e}}}}^H\left(k\right){\underline{\mathbf{H}}}^H\left(k\right)\underline{\mathbf{H}}\left(k\right)\underline{\tilde{\mathbf{e}}}\left(k\right)s\left(k\right)+{\underline{\tilde{\mathbf{e}}}}^H\left(k\right){\underline{\mathbf{H}}}^H\left(k\right)\underline{\mathbf{n}}\left(k\right)\\ & =\sqrt{P\left(k\right)}D\left(k\right)s\left(k\right)+\hat{n}\left(k\right),\end{array}$$

• where D(k) is the beam-steenng gamin for the k-th subband, which can be expressed as

$$D\left(k\right)={\underline{\tilde{\mathbf{e}}}}^H\left(k\right){\underline{\mathbf{H}}}^H\left(k\right)\underline{\mathbf{H}}\left(k\right)\underline{\tilde{\mathbf{e}}}\left(k\right),$$

• n̂(k) is AWGN with zero mean and a noise variance of σ²D(k).

The received signal-to-noise ratio (SNR) for the k-th subband with beam= steering may be expressed as: $${\gamma }_{\mathit{bs}}\left(k\right)=\frac{P\left(k\right)D\left(k\right)}{{\sigma }^2},\mathrm{for}k\in \left\{1,\dots ,N_F\right\}\mathrm{.}$$

The spectral efficiency for the k-th subband may be computed based on a continuous, monotonically increasing logarithmic function for capacity, as follows: $$C_{\mathit{bs}}\left(k\right)={\log }_2\left(1+{\gamma }_{\mathit{bs}}\left(k\right)\right),\mathrm{for}k\in \left\{1,\dots ,N_F\right\}\mathrm{.}$$
The spectral efficiency is given in units of bit/second per Hertz (bps/Hz). The mean (average) spectral efficiency for the N_F subbands of the MIMO-OFDM system may then be expressed as: $${\overline{\stackrel{‾}{C}}}_{\mathit{bs}}=\frac{{\displaystyle \sum _{k=1}^{N_F}}{C}_{\mathit{bs}}\left(k\right)}{N_F}\mathrm{.}$$

Similar computations may be performed for the beam-forming technique.

For the MISO-OFDM system, the beam-steenng is also performed for each subband using a normalized steering vector obtained for that subband. The normalized steering vector for MISO may be obtained in similar manner as that described above for the normalized steering vector ẽ(k) for the principal eigenmode (i.e., using the phase of the steenng vector). The beam-steering for MISO may be performed as shown in equation (6).

If the total transmit power for all N_T transmit antennas is limited to a particular value P_total, then the beam-forming technique may provide better results than the beam-steenng technique. This is because the total transmit power may be more optimally distributed across the N_T transmit antennas based on the eigenvectors e₁(k) for the principal eigenmode. However, if the transmit power available for each transmit antenna is limited (e.g., to P_total / N_T), then the beam-steering technique would likely achieve better results than the beam-forming technique. This is because the beam-steenng technique can more fully use all of the power available for each transmit antenna.

In any case, the total transmit power P_total may be distributed across the N_T transmit antennas and the N_F subbands using various power allocation schemes. These schemes include (1) full channel inversion, (2) selective channel inversion, (3) uniform, and (4) "water-filling" or "water-pouring" power allocation schemes. For clarity, each of these schemes is specifically described below for the beam-steering technique.

If the same amount of transmit power is used for each subband, then beam-steering can result in different received SNRs for the N_F subbands. To maximize spectral efficiency, a different coding and modulation scheme may then be used for each subband depending on the SNR achieved for the subband. However, coding and modulating individually for each subband can significantly increase the complexity of both the transmitter and receiver. On the other hand, if the same coding and modulation scheme is used for all subbands, then there may be significant variation in the error rates for the N_F subbands, depending on the variation in the received SNRs.

Full channel inversion may be used to effectively "invert" the subbands such that the received SNRs for all subbands are approximately equal. The power allocation may be performed under the constraint that the total power allocated to all subbands for each transmit antenna is limited to P_ant = P_total/N_T. For full channel inversion, the amount of transmit power P(k) to allocate to each subbands may be expressed as: $$P\left(k\right)=\frac{{\alpha }_k{P}_{\mathit{total}}}{N_T{N}_F},\mathrm{for}k\in \left\{1,\dots ,N_F\right\},$$

• where α_k is a scaling factor used for the full channel inversion power allocation. The scaling factor for the k-th subband may be expressed as:

$${\alpha }_k=\frac{b}{\left|D\left(k\right)\right|},$$

• where b is a normalization factor that may be expressed as:

$$b=\frac{1}{{\displaystyle \sum _{k=1}^{N_F}}{\left|D\left(k\right)\right|}^{-1}}\mathrm{.}$$

As shown in equations (12) and (13), the total transmit power P_total is distributed unevenly across the N_F subbands based on the scaling factors α_k, for k ∈ {1, ..., N_F}, which are inversely related to the beam-steering gains D(k). The scaling factors α_k ensure that the received SNRs for all subbands are approximately equal. The received signal power P_rx(k) for each subband may be given as: $$P_{\mathit{rx}}\left(k\right)=P\left(k\right)D{\left(k\right)}^2=\frac{{\alpha }_k{P}_{\mathit{total}}D{\left(k\right)}^2}{N_T{N}_F}=\frac{b{P}_{\mathit{total}}D\left(k\right)}{N_T{N}_F},\mathrm{for}k\in \left\{1,\dots ,N_F\right\}\mathrm{.}$$
The noise power is given by σ²D(k). The signal-to-noise ratio γ(k) for subband k is then given by: $$\gamma \left(k\right)=\frac{P\left(k\right)D{\left(k\right)}^2}{{\sigma }^{2}D\left(k\right)}=\frac{P\left(k\right)D\left(k\right)}{{\sigma }^2}=\frac{{\alpha }_k{P}_{\mathit{total}}D\left(k\right)}{N_T{N}_F{\sigma }^2}=\frac{b{P}_{\mathit{total}}}{N_T{N}_F{\sigma }^2}\mathrm{.}$$
The total received signal power P_rx may then be given as: $${\bar{P}}_{\mathit{rx}}={\displaystyle \sum _{k=1}^{N_F}}P\left(k\right)D{\left(k\right)}^2=\frac{b{P}_{\mathit{total}}}{N_T{N}_F}{\displaystyle \sum _{k=1}^{N_F}}D\left(k\right)\mathrm{.}$$

The total transmit power P_total is allocated to the subbands such that they achieve equal received SNRs (i.e., the received SNR for each subband is not a function of k), as shown in equation (16). This then enables the use of a common coding and modulation scheme for all subbands while satisfying the per-antenna power constraint.

To achieve approximately equal received SNRs for all N_F subbands, the full channel inversion scheme allocates more transmit power to poorer subbands with low gains, Because the per-antenna power is constrained to P_total/N_T, the better subbands with higher gains are allocated less transmit power. This can result in a reduction in the overall spectral efficiency of the system. However, the full channel inversion may simplify the processing at the receiver since the overall channel is effectively flat and equalization of the received signal may not be required.

The selective channel inversion scheme distributes the total transmit power P_total such that the subbands selected for use achieve approximately equal received SNR. This may be performed by first selecting all or only a subset of the N_F subbands for use for data transmission. The channel selection may result in the elimination of poor subbands with low SNR that fall below a particular threshold. This threshold may be selected to maximize spectral efficiency, as described below. The total transmit power P_total is then distributed across only the selected subbands and such that their received SNRs are approximately equal.

The scaling factors α̃_k used for power allocation by the selective channel inversion scheme may be expressed as: $${\tilde{\alpha }}_k=\{\begin{array}{ll}\frac{\tilde{b}}{{\left|D\left(k\right)\right|}^{-1}}& ,\mathrm{if}\left|D\left(k\right)\right|>\rho L_{\mathit{avg}}\\ 0& ,\mathrm{otherwise},\end{array}$$

• where ρ is a value used to set the threshold, L_avg is the average gain, and b̃ is a normalization factor. The normalization factor b̃ is similar to b in equation (14) but is computed over only the selected subbands, and may be expressed as:

$$\tilde{b}=\frac{1}{{\displaystyle \sum _{\left|D\left(k\right)\right|\ge \rho L_{\mathit{avg}}}}{\left|D\left(k\right)\right|}^{-1}}\mathrm{.}$$

$$L_{\mathit{avg}}=\frac{{\displaystyle \sum _{k=1}^{N_F}}\left|D\left(k\right)\right|}{N_F}\mathrm{.}$$

As shown in equation (17), a given subband is selected for use if its beam steenng gain is greater than or equal to the threshold (i.e., |D(k)| ≥ ρL_avg). Since no transmit power is allocated to poor subbands with gains below the threshold, higher spectral efficiency may be attained. For the subbands selected for use, the total transmit power P_total is distributed to these subbands based on their scaling factors α̃_k, similar to that shown in equation (15), such that the received signal power for each selected subband is given as b̃P_totalD(k)/N_TN_F and all selected subbands have approximately equal received SNR.

The threshold used to select subbands may be set based on various criteria. The threshold that maximizes spectral efficiency may be determined as follows. Initially, the gains D(k) for all N_F subbands are ranked and placed in descending order in a list G(λ), for λ ∈ {1, ..., N_F}, such that G(1) = max{D(k)} and G(N_F) = min{D(k)}. A sequence B(λ) is then defined as follows: $$B\left(\mathrm{\lambda }\right)={\left({\displaystyle \sum _{i=1}^{\mathrm{\lambda }}}\left(G{\left(i\right)}^{-1}\right)\right)}^{-1},\mathrm{for\; \lambda }\in \left\{1,\dots ,N_F\right\}\mathrm{.}$$

• B(λ) is the list of b̃ if the best λ, subbands are used.

The received SNR on all the selected subbands, which results when the λ best subbands are selected for use, is given as: $$\hat{\mathit{\gamma }}\left(\mathrm{\lambda }\right)=\frac{B\left(\mathrm{\lambda }\right)P_{\mathit{total}}}{{\sigma }^2{N}_T}\mathrm{.}$$
For equation (21), the total transmit power P_total is allocated among the λ best subbands such that they achieve equal received SNRs.

If the λ best subbands are selected for use, then the total spectral efficiency for these subbands is given as: $$C\left(\mathrm{\lambda }\right)=\mathrm{\lambda }{\log }_2\left(1+\hat{\gamma }\left(\mathrm{\lambda }\right)\right)\mathrm{.}$$

The spectral efficiency C(λ) may be computed for each value of λ, for λ∈ {1, ..., N_F}, and stored in an array. After all N_F values of C(λ) have been computed for the N_F possible combmations of selected subbands, the array of spectral efficiencies is traversed and the largest value of C(λ) is determined. The value of λ, λ_max, corresponding to the largest C(λ) is then the number of subbands that results in the maximum spectral efficiency for the channel conditions being evaluated.

The value ρ may then be computed as: $$\rho =\frac{G\left({\mathrm{\lambda }}_{\max }\right)}{L_{\mathit{avg}}},$$

• where L_avg is determined as shown in equation (19). The threshold ρL_avg can thus be set equal to D(λ_max), which is the gain of the worst subband in the group of subbands that maximizes spectral efficiency. The threshold used for channel selection may also be set based on some other criterion.

The received SNRs for all selected subbands can be made approximately equal by distributing the total transmit power P_total non-uniformly across these subbands. The equal received SNRs would then allow for the use of a single data rate and a common coding and modulation scheme for all selected subbands, which would greatly reduce complexity for both the transmitter and receiver.

The full and selective channel inversion schemes are described in further detail in U S. Patent Application Serial No. 09/860,274, filed May 17, 2001, Serial No 09/881,610, filed June 14, 2001, and Serial No. 09/892,379, filed June 26, 2001, all three entitled "Method and Apparatus for Processing Data for Transmission in a MultiChannel Communication System Using Selective Channel Inversion," assigned to the assignee of the present application and incorporated herein by reference.

The water-filling scheme may be used to optimally distribute the total transmit power across the subbands such that the overall spectral efficiency is maximized, under the constraint that the total transmit power is limited to P_total. The water-filling scheme distributes power to the N_F subbands such that the subbands with increasingly higher gains receive increasingly greater fractions of the total transmit power. The transmit power allocated to a given subband is determined by the subband's received SNR, which is dependent on the subband's gain, as shown in equation (9) for the beam-steenng technique. The water-filling scheme may allocate zero transmit power to subbands with sufficiently poor received SNRs.

The procedure for performing water-filling is known in the art and not described herein One reference that describes water-filling is "Information Theory and Reliable Communication," by Robert G Gallager, John Wiley and Sons, 1968, which is incorporated herein by reference The result of the water-filling is a specific transmit power allocation P_w(k) for each of the N_F subbands. The water-filling power allocation is performed such that the following condition is satisfied: $$P_{\mathit{total}}={\displaystyle \sum _{k=1}^{N_F}}{P}_w\left(k\right)\mathrm{.}$$

Based on the allocated transmit powers of P_w(k) for k = {1,..., N_F}, where P_w(k) may be zero for one or more subbands, the received SNR for each subband may be expressed as: $${\gamma }_w\left(k\right)=\frac{P_w\left(k\right)D\left(k\right)}{{\sigma }^2},\mathrm{for}k\in \left\{1,\dots ,N_F\right\}\mathrm{.}$$
The spectral efficiency C for each subband may then be computed as shown in equation (10), and the average spectral efficiency for all N_F subbands may be computed as shown in equation (11).

The water-filling power allocation typically results in different received SNRs for the subbands that have been allocated non-zero transmit powers. Different coding and modulation schemes may then be used for the selected subbands based on their received SNRs.

The uniform power allocation scheme distributes the total transmit power P_tolal uniformly across all N_F subbands. The transmit power P_u(k) allocated to each subband may be expressed as: $$P_u\left(k\right)=\frac{P_{\mathit{total}}}{N_T{N}_F},\mathrm{for}k\in \left\{1,\dots ,N_F\right\}\mathrm{.}$$

The uniform power allocation may also result in different received SNRs for the N_F subbands. Different coding and modulation schemes may then be used for these subbands based on their received SNRs. If the MIMO system has a large diversity order, then the full and selective channel inversion schemes offer little advantage over the uniform power scheme. If the MIMO system has a large diversity order, then the N_F largest eigenvalues for the N_F subbands are not likely to vary widely. In that case, the performance of the full and selective channel inversion schemes would be similar to that of the uniform power scheme.

The total transmit power may also be allocated to the subbands based on some other power allocation schemes, and this is within the scope of the invention.

Simulations were performed for (1) the beam-steenng technique with three different power allocation schemes (full channel inversion, selective channel inversion, and uniform) and (2) the beam-forming technique with uniform power allocation. When the transmit power available for each transmit antenna is limited (e.g., to P_total / N_T), the beam-steering technique provides approximately 2.5dB improvement in performance over the beam-forming technique. This significant improvement can be attributed to the fact that all of the available power is used by the beam-steering technique, which is not the case with the beam-forming technique. At a sufficiently low received SNR (which is -1 dB for the specific system configuration used in the simulations), the beam-steering technique can provide improved performance over a technique that transmits data using all of the eigenmodes and allocates the total transmit power uniformly across these eigenmodes. This is because at sufficiently low received SNRs, only a few eigenmodes are "active", and better performance may be achieved by allocating the total transmit power to the best eigenmodes. For the beam=steering technique, selective channel inversion performs better than full channel inversion at low received SNRs and when the estimates of the MIMO channel are noisy. The simulations suggest that, at low received SNRs, beam steering with selective channel inversion is a better choice for use than other MIMO transmission schemes

FIG. 2 is a block diagram of an embodiment of a transmitter system 210 and a receiver system 250 in a MIMO-OFDM system 200.

At transmitter system 210, traffic data (i.e., information bits) from a data source 212 is provided to a transmit (TX) data processor 214, which codes, interleaves, and modulates the data to provide modulation symbols. A TX spatial processor 220 further processes the modulation symbols to provide preconditioned symbols, which are then multiplexed with pilot symbols and provided to N_T OFDM modulators (MOD) 222a through 222t, one modulator for each transmit antenna. Each OFDM modulator 222 processes a respective stream of preconditioned symbols to generate a modulated signal, which is then transmitted from a respective antenna 224.

At receiver system 250, the modulated signals transmitted from the N_T antennas 224a through 224t are received by N_R antennas 252a through 252r. The received signal from each antenna 252 is provided to a respective OFDM demodulator (DEMOD) 254. Each OFDM demodulator 254 conditions (e.g., filters, amplifies, and frequency downconverts) the received signal, digitizes the conditioned signal to provide samples, and further processes the samples to provide a stream of received symbols. An RX spatial processor 260 then processes the N_R received symbol streams to provide recovered symbols, which are estimates of the modulation symbols transmitted by the transmitter system.

The processing for the reverse path from the receiver system to the transmitter system may be similar to, or different from, the processing for the forward path. The reverse path may be used to send back channel state information (CSI) from the receiver system to the transmitter system. The CSI is used at the transmitter system to (1) select the proper data rate(s) and coding and modulation scheme(s) to use for data transmission, (2) perform beam-steering or beam-forming, and (3) allocate the total transmit power to the subbands. The CSI may be provided in various forms. For example, to perform bearn-steering, the CSI may include N_T phases for the N_T transmit antennas for each subband selected for use.

Controllers 230 and 270 direct the operation at the transmitter and receiver systems, respectively. Memories 232 and 272 provide storage for program codes and data used by controllers 230 and 270, respectively.

The block diagram of the transmitter and receiver systems in a MISO-OFDM system would be similar to that shown in FIG. 2. However, the receiver system would include only one receive antenna and no RX spatial processor 260.

FIG. 3 is a block diagram of a transmitter unit 300, which is an embodiment of the transmitter portion of transmitter system 210 in FIG. 2.

Within TX data processor 214, an encoder 312 receives and codes the traffic data (i.e., the formation bits) in accordance with one or more coding schemes to provide coded bits. A channel interleaver 314 then interleaves the coded bits based on one or more interleaving schemes to provide time, spatial, and/or frequency diversity. A symbol mapping element 316 then maps the interleaved data in accordance with one or more modulation schemes (e.g., QPSK, M-PSK, M-QAM, and so on) to provide modulation symbols.

The coding and modulation for the subbands may be performed in various manners. If the received SNRs for the subbands are approximately equal at the receiver system (e.g, with full or selective channel inversion), then a common coding and modulation scheme may be used for all subbands used for data transmission. If the received SNRs are different, then a separate coding and modulation scheme may be used for each subband (or each group of subbands with approximately equal received SNRs). Convolutional, trellis, and Turbo coding may be used to code the data.

Within TX spatial processor 220, estimates of the impulse response of the MIMO channel are provided to a fast Fourier transform (FFT) unit 322 as a sequence of matrices of time-domain samples, Ĥ(n). FFT unit 322 then performs an FFT on each set of N_F matrices Ĥ(n) to provide a corresponding set of N_F estimated channel frequency response matrices, Ĥ(k) for k ∈ {1, ..., N_F}.

A unit 324 then performs eigenvalue decomposition on each matrix Ĥ(k) to provide the unitary matrix E(k) and the diagonal matrix D(k), as described above. A set of gains D(k) is then computed based on the matrices Ĥ(k) and the steenng vectors, which may be ẽ(k) or e₁(k), for k ∈ {1, ..., N_F}. The gains D(k) are provided to a power allocation unit 330 and the steering vectors are provided to a beam-steering/formmg unit 350.

Power allocation unit 330 distributes the total transmit power P_total to the subbands using any one of the power allocation schemes described above. This results in power allocations of P(k), for k ∈ {1, .., N_F}, for the N_F subbands, where P(k) may be zero for one or more subbands Power allocation unit 330 then provides scaling values $\sqrt{P\left(k\right)}$ for the subbands to a signal scaling unit 340.

The block diagram of the transmitter unit in a MISO-OFDM system would be similar to that shown in FIG. 3. However, the steering vector for each subband is derived based on a channel response vector ĥ(k) instead of the channel response matnx Ĥ(k).

FIG. 4 is a block diagram of an embodiment of a signal scaling unit 340a, a beam-steering unit 350a, and a multiplexer 360a within transmitter unit 300, which are designed to perform beam-steering Within signal scaling unit 340a, the modulation symbols s(k) are demultiplexed by a demultiplexer 440 into (up to) N_F substreams, one substream for each subband to be used for data transmission. Each symbol substream s_k is provided to a respective multiplier 442.

Each multiplier 442 performs signal scaling for an associated subband based on the scaling value $\sqrt{P\left(k\right)}$ provided for that subband. In particular, each multiplier 442 scales each modulation symbol m its substream with its scaling value $\sqrt{P\left(k\right)}$ to provide a corresponding scaled modulation symbol. The signal scaling for each modulation symbol may be expressed as_: $${\tilde{s}}_k=s_k\sqrt{P\left(k\right)}\mathrm{.}$$
The sealing value $\sqrt{P\left(k\right)}$ for each multiplier 442 is determined by the transmit power P(k) allocated to the associated subband Each substream of scaled modulation symbols s̃_k is then provided to a respective beam-steering unit 450.

Each beam-steering unit 450 performs beam-steering for an associated subband and also receives the normalized steering vector ẽ(k) for that subband. Within each unit 450, the scaled modulation symbols s̃_k are provided to N_T multipliers 452a through 452t, one multiplier for each transmit antenna. Each multiplier 452 also receives a respective element ẽ_i(k) of the normalized steering vector ẽ(k), multiplies each scaled modulation symbol in the substream with the element ẽ_i(k), and provides a preconditioned symbol x_i(k) to a combiner 460 for the transmit antenna associated with that multiplier The preconditioning performed by beam-steering unit 450k for the k-th subband may be expressed as: $$x_i\left(k\right)={\tilde{e}}_i\left(k\right){\tilde{s}}_k,\mathrm{for}i\in \left\{1,\dots ,N_T\right\}\mathrm{.}$$
Each beam-steenng unit 450 provides N_T preconditioned symbols, x_i(k) for i ∈ {1,.., N_T}, to N_T combiners 460a through 460t for the N_T transmit antennas.

The signal scaling and preconditioning may also be combined or performed in a different order than that described above.

Each combiner 460 receives up to N_F preconditioned symbols, x_i(k) for k ∈ {1, ..., N_F}, from up to N_F beam-steenng units 450 for the up to N_F subbands used for data transmission. Each combiner 460 may also multiplex pilot symbols with the preconditioned symbols in one or more subbands using time division multiplexing, coding division multiplexing, and/or frequency division multiplexing. The pilot symbols may be used at the receiver to estimate the MIMO channel Each combiner 460 provides a stream of preconditioned symbols to a respective OFDM modulator 222.

Within each OFDM modulator 222, an IFFT umt 472 receives the stream of preconditioned symbols and forms a preconditioned symbol vector x_i (n) for each symbol period. Each such vector has N_F elements for the N_F subbands, and includes preconditioned symbols for the selected subbands and zeros for the unselected subbands (i.e , x_i (n) = [x_i (1) x_i (2) ... x_i (N_F)]. IFFT unit 472 then performs an inverse FFT on each vector to obtain a corresponding time-domain representation, which is referred to as an OHM symbol. For each OFDM symbol, a cyclic prefix generator 474 repeats a portion of the OFDM symbol to form a corresponding transmission symbol. The cyclic prefix ensures that the transmission symbol retains its orthogonal properties in the presence of multipath delay spread. A transmitter (TMTR) 476 then converts the transmission symbols into one or more analog signals and further conditions (e.g , amplifies, filters, and frequency upconverts) the analog signals to generate a modulated signal that is then transmitted from the associated antenna 224

Fig. 5 is a flow diagram of an embodiment of a process 500 for transmitting data on a single eigenmode of a multiple-input channel using beam-steering or beam-forming The multiple-input channel may be a MIMO channel in a MIMO system or a MISO channel in a MISO system. Initially, a steering vector is obtained for each of the N_F subbands (step 512). The steering vector for each subband may be the eigenvector e₁(k) for the eigenmode of that subband (for beam-forming) or the normalized steering vector ẽ(k) derived based on the eigenvector e₁(k) (for beam-steering). For the MIMO system, the eigenvectors for the subbands may be obtained by performing eigenvalue decomposition on the matrices Ĥ(k), for k ∈ {1,..., N_F}, as described above. For the MISO system, there is only one eigenmode and one steering vector for each subband. Each steering vector includes N_T elements for the N_T transmit antennas. The gain D(k) for each subband provided by its steering vector is then determined (e g., as shown in equation (8) for beam-steering) (step 514).

The total transmit power P_total is allocated to the subbands using any one of the power allocation schemes described above (e.g., full channel aversion, selective channel inversion, uniform, or water-filling) (step 516) The gains for the subbands may be used to perform the power allocation. All or only a subset of the N_F subbands may be selected for use for data transmission by the power allocation. A scaling value $\sqrt{P\left(k\right)}$ is then obtained for each selected subband based on its allocated power (step 518).

Data to be transmitted is coded and modulated based on one or more coding and modulation schemes to obtain modulation symbols (step 520) A common coding and modulation scheme may be used if the received SNRs for the subbands are approximately equal. In general, the particular coding and modulation scheme to use for each subband is dependent on the received SNR achieved by that subband.

The modulation symbols to be transmitted on each subband are then scaled with the subband's scaling value (step 522). The scaled modulation symbols for each subband are then preconditioned with the subband's steering vector (step 524). The preconditioning achieves beam-steering or beam-forming for the subband, depending on whether ẽ(k) or e₁ (k) is used as the steering vector. For each subband selected for use, one vector of N_T preconditioned symbols is generated for each scaled modulation symbol, and these N_T preconditioned symbols are to be transmitted on that subband from the N_T transmit antennas.

A stream of preconditioned symbols is then formed for each transmit antenna by multiplexing the outputs of the preconditioning for the selected subbands (step 526). Each preconditioned symbol stream is further processed (e.g., OFDM modulated) to provide a modulated signal for transmission from a respective transmit antenna (step 528).

For clarity, specific embodiments have been described above. Variations to these embodiments and other embodiments may also be derived based on the teachings described herein. For example, a set of subbands may be selected for use for data transmission based on one or more criteria, independent of the scheme used to allocate transmit power to the subbands. As another example, the gains D(k) and steering vectors may be derived by the receiver system and provided to the transmitter system as part of the CSI. The processing for MIMO and MIMO-OFDM systems is described in further detail in U.S. Patent Application Serial No. 09/993,087, entitled "Multiple-Access Multiple-Input Multiple-Output (MIMO) Communication System," filed November 6, 2001, assigned to the assignee of the present application and incorporated herein by reference.

For clarity, the techniques for performing beam-steering and beam-forming have been described specifically for a MIMO-OFDM system. These techniques may also be used for a MIMO system that does not employ OFDM. The processing to achieve beam-steenng or beam-forming for each subband may be performed as described above. However, the processing by modulators 222 would be dependent on the particular modulation/transmission scheme selected for use.

The techniques described herein may be implemented by various means. For example, these techniques may be implemented in hardware, software, or a combination thereof. For a hardware implementation, the elements used to implement any one or a combination of the techniques (e.g., TX spatial processor 220) may be implemented within one or more application specific integrated circuits (ASICs), digital signal processors (DSPs), digital signal processing devices (DSPDs), programmable logic devices (PLDs), field programmable gate arrays (FPGAs), processors, controllers, micro-controllers, microprocessors, other electronic units designed to perform the functions described herein, or a combination thereof.

For a software implementation, the techniques described herein may be implemented with modules (e.g., procedures, functions, and so on) that perform the functions described herein. The software codes may be stored in a memory unit (e.g., memory unit 232 in FIG. 2) and executed by a processor (e.g., controller 230). The memory unit may be implemented within the processor or external to the processor, in which case it can be communicatively coupled to the processor via various means as is known in the art.

Headings are included herein for reference and to aid in locating certain sections. These headings are not intended to limit the scope of the concepts descnbed therein under, and these concepts may have applicability in other sections throughout the entire specification.

The previous description of the disclosed embodiments is provided to enable any person skilled in the art to make or use the present invention. Various modifications to these embodiments will be readily apparent to those skilled in the art, and the generic principles defined herein may be applied to other embodiments without departing from the scope of the claims. Thus, the present invention is not intended to be limited to the embodiments shown herein.