CROSS-REFERENCE TO RELATED APPLICATIONS
[0001] This application is based upon and claims the benefit of priority from the prior
Japanese Patent Application No. 2005-029493, filed on February 4, 2005, the entire
contents of which are incorporated herein by reference.
BACKGROUND OF THE INVENTION
[0002] The present invention relates to an apparatus for communications, in particular,
an apparatus for communications preferably using for a demodulation apparatus demodulating
signals quadrature modulated.
[0003] The quadrature amplitude modulation/demodulation is known as modulation/demodulation
technology, which is used for signal conveyance in the communications such as CATV
(cable television), satellite communications, multi-channel radio communications.
Fig. 9 is a block diagram showing a configuration of a demodulator for demodulating
quadrature amplitude modulation wave (modulated signal) in the prior art. As shown
in Fig. 9, the demodulator is composed of an RF unit 11, a MIX unit 12, A/D converters
13a, 13b, a rotator (AFC: Auto Frequency Control) 14, interpolators 91a, 91b, a rotator
(CR: Carrier Recovery) 16, low-pass filters (LPFs) 17a, 17b, a clock (CLK) phase error
operator 18, and a decoder 19.
[0004] A modulated signal input from a cable or an antenna is performed for frequency conversion
in the RF unit 11, so that its frequency is converted into the frequency possibly
to demodulate. Subsequently, at the MIX unit 12, the signal is quadrature demodulated
into base band signals as I/Q axes (hereafter, I axis is called as "I channel", Q
axis is called as "Q channel"), which are digitized by the A/D converters 13a, 13b.
[0005] That is, the modulated wave, which is frequency converted by the RF unit 11, is quadrature
demodulated into the signals corresponding to the phase axes of I channel as in-phase
and Q channel as quadrature-phase, and analog formatted I channel and Q channel signals
are output. The analog formatted I channel signal is converted into a digital formatted
I channel signal by the A/D converter 13a, and similarly, the analog formatted Q channel
signal is converted into a digital formatted Q channel signal by the A/D converter
13b.
[0006] In addition, the A/D converters 13a, 13b may be arranged at different locations from
those in Fig. 9 depending upon the processing method. For example, if they are arranged
in front of the quadrature demodulation i.e. MIX unit 12, the quadrature demodulation
within the MIX unit 12 will become a digital process.
[0007] The digital data of I/Q channels respectively output from the A/D converters 13a,
13b are synchronized with a carrier. This process is performed by shifting frequency
in the rotators 14 and 16; roughly adjusting (coarse adjustment) in the rotator (AFC)
14 and perfectly synchronizing (fine adjustment) in the rotator (CR) 16. However,
if the deviation of frequency is small, the process in the rotator (AFC) 14 may be
omitted.
[0008] Since sampling in the A/D converters 13a and 13b is performed by using clock signal
Fclk1 asynchronized to clock signal Fclk2 which is synchronized with the clock signal
of the transmission side, in order to restore digital data from the demodulated base
band signals of I/Q channels, it is necessary to provide synchronization for sampling
timing, which is generally called clock synchronization. Fig. 9 illustrates a configuration
which is not for synchronization between the clock signal Fclk1 of sampling timing
and data, but for synchronization through adjusting the phase of data, and which creates
a signal of a phase position obtaining data thereof, with an interpolation process
through the interpolators 91a and 91b in order to synchronize the phase of the data.
The created data become those synchronizing with the timing Fclk2, which is synchronized
with the clock signal of the transmission side.
[0009] The base band signals of I/Q channels phase-adjusted by the interpolators 91a and
91b are finally synchronized with the carrier by the rotator (CR) 16; wave-shaping
is carried out through low-pass filters (LPFs) 17a and 17b, which are called roll-off
filters, and thereby signals are created. In general, because of processing such as
error correcting and framing on the digital data, they will be decoded by the decoder
(DEC) 19 as necessary.
[0010] The interpolating operation through the interpolators 91a and 91b shown in Fig. 9
will be described, referring to Fig. 10. In Fig. 10, DTI is input data, which is a
sine wave in the figure for easy understanding of it. STA1 to STA4 shown by broken
lines are sampling timings for the input data DTI based on the clock signal Fclk1,
and STB1 to STB4 shown by chain-dot lines are original sampling timings based on the
clock signal Fclk2.
[0011] In order to obtain signals at positions where original sampling timings are present
from the signals sampled by the A/D converters 13a and 13b, original data sequence
ID1 to ID4 at desirable sampling timings STB1 to STB4 are calculated and produced,
from each point of the output data sequence SD1 to SD4 of the A/D converters 13a and
13b at sampling timings STA1 to STA4. Information relating to shifting phases Δt1
to Δt4 are output as phase error information PSI from the clock phase error operator
18 shown in Fig. 9.
[0012] Fig. 11 is a block diagram showing a configuration of the clock phase error operator
18. The clock phase error operator 18 comprises a clock phase error extractor 111,
a phase comparator (PD) 112, a loop filter (LPF) 113, and an oscillator (NCO) 114.
[0013] A phase error is extracted from input I/Q channel data through the clock phase error
extractor 111. The extracted phase error is compared with output timing of the oscillator
114 by the phase comparator 112. With processing of the clock phase error extractor
111 and the phase comparator 112, the phase error Δt is calculated.
[0014] Then, the comparison result at the phase comparator 12 is integrated (smoothed out)
by the loop filter 113, whereby the oscillator 114 is controlled on the basis of the
integration result. That is, a PLL (Phase Locked Loop) is composed of the phase comparator
112, the loop filter 113, and the oscillator 114. Though the oscillator 114 outputs
the timing of clock signal Fclk2, actually it will inform the interpolators 91a and
91b about the phase difference from the clock signal Fclk1 as phase error information.
[0015] In US Patent No. 5872818, detecting a symbol timing error from an output signal of
a matching filter through a timing error detector and changing tap coefficients of
the matching filter based on the detected timing error information are described.
SUMMARY OF THE INVENTION
[0016] An object of the present invention is to provide an apparatus for communications
that can suitably control a band characteristic of interpolators on the basis of a
receiving signal.
[0017] An apparatus for communications according to the present invention includes a front
processor for outputting a digital formatted base band signal by quadrature demodulating
a modulated signal, an interpolation processor for interpolating to adjust a phase
error in accordance with sampling timing on the basis of the base band signal output
from the front processor and for creating and outputting a base band signal synchronized
with the sampling timing, and a back processor for wave-shaping the base band signal
output from the interpolation processor and for outputting. The interpolation processor
includes an FIR filter which controls tap coefficients supplying to the FIR filter
on the basis of the modulated signal input therein. The invention is also concerned
with corresponding methods.
[0018] According to the configuration as described above, it is possible to control a band
characteristic of the FIR filter by supplying tap coefficients on the basis of the
receiving modulated signal to the FIR filter.
BRIEF DESCRIPTION OF THE DRAWINGS
[0019] For a better understanding of the invention, embodiments of it will now be described,
by way of example, with reference to the accompanying drawings, in which:
Fig. 1 is a block diagram showing an example of a configuration of a demodulator according
to a first embodiment;
Figs. 2A and 2B are diagrams showing an example of a configuration of an interpolator;
Figs. 3A and 3B are diagrams showing examples of changing bandwidth of the interpolator;
Fig. 4 is a block diagram showing an example of a configuration of a demodulator according
to a second embodiment;
Fig. 5 is a block diagram showing an example of a configuration of a demodulator according
to a third embodiment;
Fig. 6 is a block diagram showing a configuration of a controller according to the
third embodiment;
Fig. 7 is a block diagram showing an example of a configuration of a demodulator according
to a fourth embodiment;
Fig. 8 is a block diagram showing a configuration of a controller according to the
fourth embodiment;
Fig. 9 is a block diagram showing a configuration of a demodulator in the prior art;
Fig. 10 is a diagram to explain an interpolating action through an interpolator;
Fig. 11 is a block diagram showing a configuration of a clock phase error operator;
Figs. 12A and 12B are diagrams showing a configuration of an interpolator within a
demodulator in the prior art;
Fig. 13 is a diagram to explain a filter band of an interpolator;
Fig. 14 is a conceptual view of an interpolator within a demodulator in the prior
art; and
Fig. 15 is a diagram to explain influence of an adjacent wave.
[0020] Each of the interpolators shown in Fig. 9 is composed of a FIR (finite impulse response)
filter 121 shown in Fig. 12A and a coefficient operator (coefficient memory) 127 shown
in Fig. 12B which outputs coefficients (tap coefficients) Ci (i is a subscript equal
to an integer between minus 4 and plus 4) to a FIR filter 121.
[0021] As shown in Fig. 12A, the FIR filter 121 is composed of a group of registers 123
which is composed of a plurality of registers 122 in cascade connection and to which
input signal data IDT are input; multipliers 124 each of which multiplies the output
of each register 122 (or input signal data IDT) by each tap coefficient Ci; and an
adder 125 which adds multiplying results of each multiplier 124 and outputs it as
output signal data ODT. In Fig. 12A, a FIR filter 121 having nine taps is illustrated
as an example.
[0022] The FIR filter 121, which holds input signal data IDT during a plurality of time,
treats each item of data as an impulse, and changes the phase by controlling the tap
coefficients Ci multiplied by each impulse. In other words, with appropriate control
of the tap coefficients Ci, the FIR filter 121 can output a signal which suitably
shifts the phase for the input signal data IDT as the output signal data ODT. These
coefficients Ci can be calculated from the impulse response waveform.
[0023] That is, the FIR filter 121 has an impulse response as a filter limiting a band (for
example, low-pass filter: LPF), and outputs a value derived from a timing position
of clock signal Fclk1 to a timing position of clock signal Fclk2 of the impulse response,
as coefficients Ci through the coefficient operator 127. This shows that the coefficients
Ci of the FIR filter 121 change from moment to moment corresponding to the change
of the phase error information PSI input to the interpolators.
[0024] The coefficient operator 127 outputs the tap coefficients Ci according to the phase
error information PSI which is input. The coefficient operator 127 has previously
stored a table including the tap coefficients Ci corresponding to the amount of the
phase errors Δt, on the basis of the phase errors indicated by the phase error information
PSI, and reads and outputs tap coefficients Ci corresponding to the phase errors.
In addition, the coefficient operator 127 may be configured to calculate the tap coefficients
Ci with calculation on an as-needed basis of the input phase error information PSI.
[0025] There follows a description of the operation of the interpolators (FIR filter 121)
in a frequency domain, in other words, the operation as a filter. Fig. 13 is a diagram
to explain a filter band of the interpolator. In Fig. 13, the interpolator operates
as a filter limiting band, and herein operates as a low-pass filter (LPF).
[0026] In Fig. 13, F1 is a band of a base band signal of I channel or Q channel inputting
to the interpolator, and BN1 is a band of the interpolator (pass band as filter).
Further, Fclk1 and Fclk2 respectively illustrate the frequency of the sampling clock
in the A/D converter and the clock frequency of the transmission side (actually desirable
sampling timing).
[0027] The filter characteristic of the interpolator is preferably a conveyance characteristic
without affecting the base band signal of the input I channel or Q channel, as shown
in Fig. 13. The band BN1 of the interpolator generally as a wider band characteristic
than that of the band F1 of the base band signal. With this operation, it is possible
to process the interpolating operation.
[0028] Though the interpolator operates as a low-pass filter, it also operates as a modulator
(modulating operation), because the tap coefficients Ci change from moment to moment
as described above. Fig. 14 is a conceptual view to explain the modulating operation
through the interpolator (FIR filter 121). In Fig. 14, blocks having functions being
identical to those of the blocks shown in Fig. 9 are indicated the same numbers/characters,
so that the description will not be repeated.
[0029] As shown in multipliers 141a, 141b in Fig. 14, the interpolators perform not only
the operation as a low-pass filter but also the operation as the multiplying operation
at the frequency (Fclk1 - Fclk2) for the base band signal of input I/Q channels. This
operation is, as shown in Fig. 15, the same operation of modulating the input signal
with the frequency (Fclk1 - Fclk2) in the frequency axis.
[0030] As shown in Fig. 15, it is desirable for the signal input to the interpolator to
be only the base band signal (band F1) of the receiving I/Q channels. However, in
many cases, a signal (band F2) of an adjacent channel remains. The signal of the adjacent
channel means a signal being adjacent to the waveform of the desirable channel when
a plurality of waveforms are arranged in the received highfrequency signal, because,
in many cases, multi-channel signals are transmitted, such as in CATV, satellite communications,
and multi-radio communications; such signals from the adjacent channels can become
an interference source in the waveform of the desirable channel.
[0031] This signal of the adjacent channel is also modulated by the interpolator. Consequently,
as shown in Fig. 15, signals modulated by the interpolator appear in bands F2A and
F2B which are lower and upper frequencies of the signal band F2 of the adjacent channel.
However, since a band limitation is also simultaneously effected by the operation
as the low-pass filter actually, while the components of higher frequency (the components
of the band F2 and F2B) are cut off from the right signal part of the adjacent channel
being out of the pass band BN1, the components of the modulated signal (the components
of the band F2A) relating to the adjacent channel within the pass band BN1 are not
cut off, and are transferred to the latter stage without being processed, and are
likely to become a noise source. The components transferred to the latter stage among
those of the adjacent channel illustrate as the hatching area NS in Fig. 15.
[0032] It is considered that the interference owing to the signal components of this hatching
area NS affects the base band signal of I/Q channels of the desirable receiving channel.
In general, since the signal components of the hatched area NS are removed by roll-off
filters (LPFs) 17a and 17b positioned at the latter stage shown in Fig. 9, it is considered
that the signal components of the hatched area NS do not affect the desirable receiving
channel finally. However, the interference components affect the dynamic range until
the input terminals of the roll-off filters (LPFs) 17a and 17b, which causes a reduction
in quality.
[0033] In particular, if the level of the signal relating to the adjacent channel is very
large, the signal level will exceed the possible range to process owing to the noise
of the hatching area NS. This will become a serious problem such that performing the
process in the latter stage is impossible.
[0034] In order to solve the problem described above, a method will be considered which
makes the band width BN1 of the interpolator narrow, of the same order of that of
the roll-off filters 17a and 17b in the latter stage. However, when the band width
BN1 is simply made narrow without study, the following problem occurs. At the time
of performing a process in the interpolator, frequency shift components exist in the
input signal owing to imperfect synchronization of the carrier with only coarse adjustment
through the rotator (AFC) 14. If the frequency shift components are cut off through
the interpolator by making the bandwidth BN1 narrow, a deterioration of the characteristic
will occur.
[0035] Accordingly, the band width of the interpolator should be determined in an optimal
manner, with consideration of the effect of the adjacent channel and the attenuation
of the self-wave owing to carrier shift. Therefore, a demodulator applied an apparatus
for communications according to embodiments of the present invention is a modulator
which can suitably control the band width of the interpolator corresponding to an
input modulated signal.
[0036] Initially, a first embodiment according to the present invention will be described.
[0037] Fig. 1 is a block diagram showing an example of a configuration of a demodulator
10 provided in the apparatus for communications representing the first embodiment
of the present invention.
[0038] The demodulator 10 includes an RF (frequency converter) unit 11, a MIX (mixer) unit
12, A/D (analog-to-digital) converters 13a and 13b, a rotator (AFC: Auto Frequency
Control) 14, interpolators 15a and 15b, a rotator (CR: Carrier Recovery) 16, low pass
filters (LPFs) 17a and 17b, a clock phase error operator 18, and a decoder 19.
[0039] The RF unit 11 performs frequency conversion on a modulated signal input therein,
and converts its frequency into a demodulatable frequency. For example, the RF unit
11 converts a demodulated frequency that is input with 1 to 2 GHz frequency into a
frequency with around 500 MHz. The MIX unit 12 quadrature demodulates (separates)
the modulated wave performed frequency conversion through the RF unit 11 into base
band signals corresponding to the phase axes of I channel and Q channel.
[0040] The A/D converters 13a and 13b digitally convert analog formatted base band signals
of I/Q channels output from the MIX unit 12 with sampling on the basis of clock signal
Fclk1. In addition, the locations of the A/D converters 13a and 13b are not limited
to this case, which may be arranged in different locations depending upon the processing
method. For example, if they are arranged in front of the MIX unit 12, the quadrature
demodulation within the MIX unit 12 will become a digital process. Further, the clock
signal Fclk1 and the clock signal Fclk2 are asynchronous, the clock signal Fclk2 is
synchronous with a clock signal of a transmission side.
[0041] The rotator (AFC) 14 and the rotator (CR) 16 make the carriers synchronous with digital
data of the I/Q channels. The rotator 14 (AFC) roughly adjusts the frequency by coarse
adjustment and the rotator (CR) 16 performs fine adjustment for perfect synchronization.
If the deviation of the frequency is small, the performance through the coarse rotator
(AFC) 14 may be omitted.
[0042] To the interpolators 15a and 15b, phase error information PSI is input from the clock
phase error operator 18, and also reception rate information SRI is input from a receiving
controller (not shown). The interpolators 15a and 15b are controlled on the basis
of the phase error information PSI, and the reception rate information SRI, etc.,
and, in order to restore digital data from the demodulated base band signals of the
I/Q channels, the interpolators 15a and 15b perform synchronism, which is generally
called as clock synchronism, for the sampling timing. Concretely, the interpolators
15a and 15b, as described by using Fig. 10, adjust the phase of data on the basis
of the data of the I/Q channels performed sampling by using the clock signal Fclkl,
and create data through calculation at the sampling timing by using the clock signal
Fclk2 synchronized with the clock signal of the transmission side.
[0043] The low-pass filters (LPFs) 17a and 17b, which are so called roll-off filters, perform
wave-shaping for the base band signals of the I/Q channels where the carriers are
synchronized by the rotator (CR) 16. The decoder 19 performs processing such as error
correction and framing performed on digital data as necessary, and also decodes and
outputs it as the data of the I/Q channels.
[0044] The clock phase error operator 18 acquires the phase difference between the clock
signal Fclk1 and the clock signal Fclk2, in other words the difference between the
actual sampling timing and the desirable sampling timing with calculation. The clock
phase error operator 18, on the basis of the calculation result, outputs the phase
difference between clock signal Fclk1 and clock signal Fclk2 as the phase error information
PSI to the interpolators 15a and 15b. The configuration of the clock phase error operator
18 is the same as shown in Fig. 11 described above, so that detailed description of
it will be omitted.
[0045] Figs. 2A and 2B are diagrams showing an example of the configuration of the interpolators 15a and
15b according to the present embodiment. Each of the interpolators 15a and 15b includes
an FIR (finite impulse response) filter 21 as shown in Fig. 2A, and a coefficient
operator 26 as shown in Fig. 2B.
[0046] The FIR filter 21, as shown in Fig. 2A, includes a group of registers 23 composed
of a plurality of registers 22 in cascade connection, multipliers 24 each of which
multiplies the output of each register 22 (or input signal data IDT) by each coefficient
(tap coefficient) Ci (i is a subscript equal to an integer in the range minus 4 to
plus 4), and an adder 25 which adds the multiplying results of each multiplier 24
and outputs them as output signal data ODT. The FIR filter 21 is here the same as
the FIR filter 121 as shown in Fig. 12A, so that detailed description of it will be
omitted. In addition, though Fig. 2A illustrates an example of the FIR filter 21 having
nine taps, it is not limited to this, the number of taps being arbitrary.
[0047] The coefficient operator 26 shown in Fig. 2B includes coefficient memories 27-1 and
27-2, and a selector 28.
[0048] The coefficient memories 27-1 and 27-2 are correspondingly provided for each state
which is indicated by control information CI, that is, one coefficient memory is provided
corresponding to one state. Each of the coefficient memories 27-1 and 27-2 has previously
stored in it a tap coefficient table in which tap coefficients Ci corresponding to
the amount of a phase error are defined, and outputs tap coefficients Ci corresponding
to a phase error indicated by input phase error information PSI to the selector 28.
[0049] The tap coefficients Ci are supplied from each of the coefficient memories 27-1 and
27-2 to the selector 28, which then selects the tap coefficients Ci from one of them
on the basis of the control information CI, and outputs them to the FIR filter 21.
[0050] That is, the coefficient operator 26 according to the present embodiment shown in
Fig. 2B selects a set of tap coefficients Ci based on the input phase error information
PSI and control information CI, and supplies the tap coefficients Ci corresponding
to the selected phase error information and control information to the FIR filter
21. In this first embodiment, the control information CI is reception rate information
SRI which is input from a receiving controller, not shown.
[0051] In addition, though Fig. 2B illustrates one example of the coefficient operator 26
including two coefficient memories 27-1 and 27-2, it is not limited to this: an arbitrary
number of coefficient memories may be provided corresponding to the control information,
etc. Further, instead of the configuration providing the plurality of coefficient
memories, a configuration in which a memory area of a coefficient memory is divided
into plural areas can be used. Furthermore, it may be configured to calculate the
tap coefficients Ci with calculation on an as-needed basis of the input phase error
information, the control information, etc.
[0052] Next, the operation will be described.
[0053] A modulated wave which is input from a cable or an antenna, after its frequency is
converted into a demodulatable frequency by the RF unit 11, is quadrature demodulated
into base band signals of I/Q channels by the MIX unit 12. Subsequently, they are
converted into the digital formatted base band signals of the I/Q channels by the
A/D converters 13a and 13b.
[0054] The base band signals of the I/Q channels digitized by the A/D converters 13a and
13b are given a rough frequency adjustment in order to synchronize with the carrier
by the rotator (AFC) 14. Then, clock synchronism is performed with the phase adjustment
in the interpolators 15a and 15b. Subsequently, perfect synchronism is performed for
the carrier through the fine adjustment of the frequency in the rotator (CR) 16. The
phase adjustment in the interpolators 15a and 15b is performed by the FIR filter 21,
the tap coefficients Ci being supplied according to the phase error information PSI
and the reception rate information SRI. The phase error information PSI is calculated
by the clock phase error operator 18 at any time as described above, and is output
on the basis of the calculation result.
[0055] The base band signals of the I/Q channels, which are finally synchronized with the
carrier in the rotator (CR) 16, are wave-shaped by the roll-off filters 17a and 17b,
and are decoded by the decoder 19, and output.
[0056] In the first embodiment, symbol rate information (symbol frequency set signal) of
a receiving channel is supplied to the interpolators 15a and 15b as the reception
rate information SRI. The bandwidth (cut-off frequency) of the interpolators 15a and
15b is controlled by changing the tap coefficients Ci of the FIR filter 21 which are
supplied by the coefficient operator 26 according to the symbol rate (symbol frequency)
indicated by the symbol rate information.
[0057] Since the bandwidth of the transmitting signal changes with the symbol rate, changing
the bandwidth of the interpolators 15a and 15b according to the symbol rate and changing
that according to the bandwidth of the transmitting signal are the same. That is,
in the first embodiment, the bandwidth of interpolators 15a and 15b is controlled
and changed according to the bandwidth of the transmitting signal.
[0058] In addition, in the case of normal reception, the symbol rate (the bandwidth of the
transmitting signal) of a receiving channel is previously known, and this information
is included in the receiving controller, not shown, etc. Accordingly, the symbol rate
information supplied to the interpolators 15a and 15b can be created and output on
the basis of the information included in the receiving controller, etc.
[0059] As described above, the bandwidth of the interpolators 15a and 15b preferably has
a conveying characteristic without affecting the received signal, and is generally
a wider band than that of the received signal. Thereby, as an example described in
Fig. 3A, the bandwidth of the interpolators 15a and 15b is wider by a constant or
given frequency amount (e.g. 5MHz) than the symbol rate (signal bandwidth).
[0060] Fig. 3A is a diagram showing a controlling example of the bandwidth of the interpolators
15a and 15b according to the present embodiment. In Fig. 3A, each of WF1, WF2 and
WF3 indicates the band where each symbol rate is SR1, SR2, and SR3; and each of BF1,
BF2, and BF3 indicates the bandwidth of interpolators 15a and 15b corresponding to
each receiving signal band WF1, WF2, and WF3.
[0061] As shown in Fig. 3A, by controlling the bandwidth of the interpolators 15a and 15b
according to the amount of the symbol rate i.e., the band of the receiving signal,
for example, the bandwidth of the interpolators can be set to the bandwidth which
is slightly wider than the band of the receiving signal and is hardly affected by
the carrier shift, corresponding to the receiving signal. In addition, as shown in
Fig. 3A, the tap coefficients Ci, by which each of the bandwidths BF1, BF2, and BF3
of the interpolators 15a and 15b is set, are previously stored in the coefficient
memories included in the coefficient operator 26 of the interpolators 15a and 15b,
so that they may appropriately be selected according to each symbol rate SR1, SR2,
and SR3 indicated by the reception rate information SRI.
[0062] In the example above, the bandwidth of the interpolators 15a and 15b is set to be
wider by the constant number of frequencies (frequency amount) than the symbol rate
(signal bandwidth). In the example shown in
Fig. 3B, it can be wider by a constant ratio (e.g. 1.3 times) in comparison to the symbol
rate (signal bandwidth). In this case also, symbol rate information (symbol frequency
set signal) of a receiving channel may be supplied to the interpolators 15a and 15b
as receive rate information; thus the bandwidth (cut-off frequency) of the interpolators
15a and 15b may be controlled by changing the tap coefficients Ci of the FIR filter
21 which are supplied by the coefficient operator 26.
[0063] Fig. 3B is a diagram showing another controlling example of the bandwidth of the
interpolators 15a and 15b according to the present embodiment. In Fig. 3B, each of
WF1, WF2 and WF3 indicates the signal band where each symbol rate is SR1, SR2, and
SR3; and each of BF11, BF12, and BF13 indicates the bandwidth of the interpolators
15a and 15b corresponding to each receiving signal band WF1, WF2, and WF3.
[0064] In this way, if the bandwidth of the interpolators 15a and 15b is set to be wider
by k times (k is an arbitrary value) in comparison to the symbol rate (signal bandwidth),
the bandwidth, for example, can be of higher accuracy and have a better characteristic
than being wider by the constant number of frequencies in the case where the dynamic
range of the symbol rate is wide.
[0065] As described above, in accordance with the first type of embodiment, the bandwidth
of the interpolators 15a and 15b (FIR filter 21) can be controlled to a suitable bandwidth
corresponding to the bandwidth of the receiving signal, by changing the tap coefficients
which are supplied to the FIR filter 21 of the interpolators 15a and 15b according
to the symbol rate information (bandwidth of receiving signal) of the receiving channel.
[0066] Next, a second embodiment according to the present invention will be described.
[0067] Fig. 4 is a block diagram showing an example of a configuration of a demodulator
40 according to the second embodiment. In Fig. 4, the blocks having functions being
identical to those of the blocks shown in Fig. 1, are indicated by the same numbers/characters,
so that duplicated description will be omitted. In Fig. 4, interpolators 41a and 41b
correspond to the interpolators 15a and 15b shown in Fig. 1, and are configured as
well as those in Figs. 2A and 2B.
[0068] The demodulator 40 according to the second embodiment shown in Fig. 4 is different
from the demodulator 10 according to the first embodiment shown in Fig. 1 with respect
to the information input to the interpolators. In the second embodiment, multi-value
information QAMI (QAM set information) is input from a receiving controller (not shown)
to the interpolators 41a and 41b of the demodulator 40, instead of the reception rate
information SRI.
[0069] The multi-value information QAMI becomes the control information CI for the interpolators
41a and 41b in the second embodiment. That is, in the second embodiment, the bandwidth
(cut-off frequency) of the interpolators 41a and 41b is changed according to a multi-value
relating to the quadrature amplitude modulation.
[0070] In this, the multi-value relating to the quadrature amplitude modulation is one of
the parameters to which error occurrence by causing the carrier shift or signal cut-off
affects. In the quadrature amplitude modulation, distortion and/or noise strength
differs depending upon the multi-value, while as for the cut-off distortion of the
self-wave with the carrier shift, its strength also differs depending upon the multi-value.
As the multi-value is greater, the strength against the influence of noise, etc. from
the adjacent channel decreases; for example, even if it sufficiently endures against
the multi-value 4 of 4PSK, there is a possibility that it does not endure at all against
the multi-value 256 of 256QAM.
[0071] For this reason, in the second embodiment, the bandwidth (cut-off frequency) of the
interpolators 41a and 41b is controlled by changing the tap coefficients Ci supplied
to the FIR filter according to the phase error information PSI and the multi-value
information QAMI. The tap coefficients Ci supplied to the FIR filter are stored into
each coefficient memory provided corresponding to each multi-value, and the tap coefficients
Ci corresponding to the multi-value indicated by the multi-value information QAMI
are supplied to the FIR filter, appropriately exchanging the selector on the basis
of the multi-value information QAMI. In addition, as described above, since, as the
multi-value is greater, the strength decreases, as the multi-value is greater, the
bandwidth (cut-off frequency) of the interpolators 41a and 41b is controlled to be
narrower. That is, the bandwidth of the interpolators 41a and 41b is changed according
to the multi-value, in accordance with the relation that the bandwidth of the interpolators
41a and 41b is inversely proportional to the multi-value.
[0072] According to the second embodiment, the bandwidth of the interpolators 41a and 41b
can be suitably controlled in a way corresponding to the multi-value of the receiving
signal, by changing the tap coefficients Ci which are supplied to the FIR filter of
the interpolators 41a and 41b according to the multi-value of the modulation system
of the receiving signal.
[0073] Next, a third embodiment according to the present invention will be described.
[0074] Fig. 5 is a block diagram showing an example of a configuration of a demodulator
50 according to the third embodiment. In Fig. 5, the blocks having functions identical
to those of the blocks shown in Fig. 1 are indicated by the same numbers/characters,
so that overlapping description will be omitted.
[0075] In Fig. 5, interpolators 51a and 51b correspond to the interpolators 15a and 15b
shown in Fig. 1, and are configured as well as those in Figs. 2A and 2B. The numeral
52a is a comparator (subtracter) where the base band signal input to the interpolator
51a and the base band signal output from the interpolator 51a are input, then the
difference between them is output. Similarly with the comparator (subtracter) 52b,
the base band signal input to the interpolator 51b and the base band signal output
from the interpolator 51b are input, then the difference of them is output. The outputs
from the comparators 52a and 52b are input to a controller 53, which outputs control
information CI on the basis of them to the interpolators 51a and 51b.
[0076] In this, if the influence of the signal of the adjacent channel on the signal of
the receiving channel is small, the influence of the self-wave reduction with the
carrier shift can be decreased by making the bandwidth (cut-off frequency) of the
interpolators wide. For this reason, the demodulator 50 shown in Fig. 5 according
to the third embodiment changes the bandwidth (cut-off frequency) of the interpolators
51a and 51b corresponding to the influence of the adjacent channel.
[0077] However, it is very difficult to directly measure the signal strength of the adjacent
channel. In the demodulator 50 according to the third embodiment, the influence of
the adjacent channel is detected by the difference between the front and back signal
levels of the interpolators 51a and 51b. That is, by detecting the difference between
the front and back signal levels (input and output) of the interpolators 51a and 51b
with the comparators 52a and 52b, the noise component with the signal of the adjacent
channel is detected. Then, on the basis of the detected result, the control information
CI is output from the controller 53, and the bandwidth (cut-off frequency) of the
interpolators 51a and 51b is controlled. In this way, the bandwidth (cut-off frequency)
of the interpolators 51a and 51b is controlled by utilizing the difference between
the front and back signal levels of the interpolators 51a and 51b.
[0078] Concretely, if the component of the adjacent channels is cutoff by the interpolators
51a and 51b, the difference between the front and back signal levels of these is little,
so that it is possible to make the bandwidth (cut-off frequency) of the interpolators
51a and 51b wide. If on the contrary the difference between the front and back signal
levels of the interpolators 51a and 51b is large, it is considered that the component
of the adjacent channels is not cutoff, so that the control is changed to make the
bandwidth (cut-off frequency) of the interpolators 51a and 51b narrow.
[0079] Fig. 6 is a block diagram showing a configuration of the controller 53 shown in Fig.
5.
[0080] In Fig. 6, the numeral 61 is an amplitude comparator which calculates the difference
between the front and back signal levels of the interpolators 51a and 51b, and which
is composed of the comparators 52a and 52b. The numeral 62 is a loop filter (LPF)
which accumulates (integrates) the outputs (comparison results of the signal levels)
of the amplitude comparator 61. The numeral 63 is an evaluation unit which compares
the output from the loop filter 62 with a previously set threshold, and outputs control
information CI according to the comparison result. The number of the previously set
threshold is arbitrary; if a plurality of thresholds are set, it is possible to finely
control the bandwidth (cut-off frequency) of the interpolators 51a and 51b.
[0081] Next, a fourth embodiment according to the present invention will be described.
[0082] Fig. 7 is a block diagram showing an example of a configuration of a demodulator
70 according to the fourth embodiment. In Fig. 7, the blocks having functions identical
to those of the blocks shown in Fig. 1, are indicated by the same numbers/characters,
so that overlapping description will be omitted.
[0083] In Fig. 7, interpolators 71a and 71b correspond to the interpolators 15a, 15b shown
in Fig. 1, and are configured as well as those in Figs. 2A and 2B. The numeral 72
is a controller which outputs control information CI on the basis of signals output
from the low-pass filters 17a and 17b.
[0084] Fig. 8 is a block diagram showing a configuration of the controller 72 shown in Fig.
7.
[0085] In Fig. 8, the numeral 81 is an error calculator which calculates a distance between
a signal point of a signal output from the low-pass filters 17a and 17b and a basic
signal point previously determined according to the modulation system, i.e. calculates
an error. The numeral 82 is a loop filter (LPF) which accumulates (integrates) outputs
from the error calculator 81, and the numeral 83 is an evaluation unit which outputs
control information CI according to the output from the loop filter 82.
[0086] That is, the controller 72 according to the fourth embodiment observes a constellation
of the output signal from the low-pass filters 17a and 17b, and outputs control information
CI according to its result, and controls the bandwidth (cut-off frequency) of the
interpolators 71a and 71b. Concretely, if the signal of the adjacent channel affects
the signal of the receiving channel, since the error of the demodulating signal, which
is calculated by the error calculator 81 where the constellation is inflated, becomes
large, the controller 72 changes the bandwidth (cut-off frequency) of the interpolators
71a and 71b narrowly.
[0087] In addition, though the first to fourth embodiments are described individually, these
may coordinate or be combined as desired in whole or part, so that the same effect
can be obtained from the coordinating as that from each embodiment.
[0088] Further, the foregoing embodiments are only one example for performing the present
invention concretely, and should not be intended to limit the technological scope.
That is, the present invention can be provided as various alternative embodiments
without departing from the technology spirit or main feature thereof.
[0089] According to the present invention, by changing the tap coefficients supplied to
the FIR filter included in the interpolation processor according to the receiving
signal, the filter band of the FIR filter can be set with a wider bandwidth than that
of the receiving signal and with hard effect from the carrier shift in each receiving
channel, and the filter bandwidth characteristic of the FIR filter can appropriately
be controlled according to the receiving signal.
1. An apparatus for communications, comprising:
a front processor (11-13) for outputting a digital formatted base band signal with
quadrature demodulation of a modulated signal;
an interpolation processor (15; 41; 51; 71) for interpolating to adjust a phase error
corresponding to sampling timing on the basis of the base band signal output from
the said front processor and for creating and outputting a base band signal synchronized
with said sampling timing; and
a back processor (17) for wave-shaping the base band signal output from the interpolation
processor and for outputting;
wherein the interpolation processor (15; 41; 51; 71) includes a FIR filter (21) and
controls tap coefficients (Ci) for supply to this FIR filter on the basis of the said
input modulated signal.
2. An apparatus according to claim 1, wherein the interpolation processor (15) further
comprises a coefficient operator (26) which outputs tap coefficients according to
the said modulated signal to the FIR filter.
3. An apparatus according to claim 2, wherein the coefficient operator (26) comprises
a plurality of coefficient memories (27) for storing the tap coefficients, and a selector
(28) which selects one coefficient memory from the plurality of coefficient memories,
according to the modulated signal.
4. An apparatus according to any preceding claim,
wherein the apparatus is adapted to change the tap coefficients supplied to the FIR
filter according to the bandwidth of the modulated signal.
5. An apparatus according to claim 4, wherein the apparatus ascertains the bandwidth
of the modulated signal on the basis of its symbol rate information.
6. An apparatus according to claim 5, wherein the said symbol rate information is a symbol
frequency set signal of the modulated signal.
7. An apparatus according to any of claims 4 to 6 and adapted to supply tap coefficients
which set a filter bandwidth wider by a constant frequency number than that of the
said modulated signal, according to the bandwidth of the modulated signal, to the
FIR filter.
8. An apparatus according to any of claims 4 to 6 and adapt to supply tap coefficients
which set a filter bandwidth wider by a constant ratio than that of the modulated
signal, according to the bandwidth of the modulated signal, to the FIR filter.
9. An apparatus according to any of claims 4 to 6, and adapted to supply tap coefficients
which set a filter bandwidth wider by k times (k being an arbitrary value greater
than or equal to 1) than that of the modulated signal, according to the bandwidth
of the modulated signal, to the FIR filter.
10. An apparatus according to any of claim 1 to 3,
wherein the apparatus changes tap coefficients supplied to the FIR filter according
to a multi-value of said modulated signal.
11. An apparatus according to claim 10, wherein the apparatus controls tap coefficients
supplied to the FIR filter so as to make the filter bandwidth of the FIR filter inversely
proportional to the multi-value of the modulated signal.
12. An apparatus according to claim 10, wherein the apparatus changes tap coefficients
supplied to said FIR filter so as to make the bandwidth of the FIR filter narrow if
the multi-value of the modulated signal is large and wide if it is small.
13. An apparatus according to any of claims 10 to 12,
wherein the apparatus ascertains the multi-value of the modulated signal by using
a multi-value set signal of the modulated signal.
14. An apparatus according to claim 1, wherein the apparatus is adapted to change tap
coefficients supplied to the FIR filter according to the signal level of a channel
adjacent to the said modulated signal.
15. An apparatus according to claim 14, wherein the apparatus is adapted to detect the
signal level of the said adjacent channel on the basis of a difference between an
input level to, and an output level from, the FIR filter.
16. An apparatus according to claim 15, further comprising:
a comparator (61) for calculating the difference between the said input level and
output level;
a loop filter (62) for integrating the outputs of the comparator; and
a control circuit (63) which outputs control information for controlling tap coefficients
supplied to the FIR filter on the basis of the output of the loop filter (62).
17. An apparatus according to claim 14, wherein the apparatus is adapted to detect the
signal level of the said adjacent channel on the basis of the constellation of a signal
output from the back processor (17).
18. An apparatus according to claim 17, further comprising:
an operator (81) for calculating the difference between a signal point of a signal
output from the back processor and a predetermined reference signal point;
a loop filter (82) for integrating outputs of the said operator; and
a control circuit (83) for outputting control information for controlling tap coefficients
to be supplied to the FIR filter on the basis of the output of the loop filter (82).